https://doi.org/10.6113/JPE.2019.19.1.11
ISSN(Print): 1598-2092 / ISSN(Online): 2093-4718
New ZVZCT Bidirectional DC-DC Converter Using Coupled Inductors
Wei Qian*, Xi Zhang†, Zhe Li*, Wenqiang Jin*,**, and Jochen Wiedemann***
†,*School of Mechanical and Power Engineering, Shanghai Jiao Tong University, Shanghai, China
**Argonne National Lab, Lemont, IL, USA
***Department of Automotive Engineering, University of Stuttgart, Stuttgart, Germany
Abstract
In this study, a novel zero voltage zero current transition (ZVZCT) bidirectional DC-DC converter is proposed by employing coupled inductors. This converter can turn the main switch on at ZVZCT and it can turn it off with zero voltage switching (ZVS) for both the boost and buck modes. These characteristics are obtained by using a simple auxiliary sub-circuit regardless of the power flow direction. In the boost mode, the auxiliary switch achieves zero current switching (ZCS) turn-on and ZVS turn off. Due to the coupling inductors, this converter can make further efficiency improvements because the resonant energy in the capacitor or inductor can be transferred to the load. The main diode operates with ZVT turn-on and ZCS turn-off in the boost mode. For the buck mode, there is a releasing circuit to conduct the currents generated by the magnetic flux leakage to the output. The auxiliary switch turns on with ZCS and it turns off with ZVT. The main diode also turns on with ZVT and turns off with ZCS. The design method and operation principles of the converter are discussed. A 500 W experimental prototype has been built and verified by experimental results.
Key words: Bidirectional, Coupled inductor, DC-DC converter, ZVZCT
Manuscript received May 9, 2018; accepted Sep. 15, 2018
Recommended for publication by Associate Editor Jongbok Baek.
†Corresponding Author: braver1980@sjtu.edu.cn Tel: +86-13918913376, Shanghai Jiao Tong University
*Sch. of Mechanical and Power Eng., Shanghai Jiao Tong Univ., China
**Argonne National Lab, USA
***Dept. of Automotive Engineering, University of Stuttgart, Germany
Ⅰ. INTRODUCTION
Recently, the research on bidirectional DC-DC converters (BDCs) has become an important topic due to its application in energy recovery and energy storage systems. DC-DC converters are widely applied as such in solar power supplies, electric vehicles, energy storage systems, fuel cell vehicles, etc. [1]-[6]. BDCs can keep storage devices healthy by controlling the discharge and charge [8]. Bi-directional converters have two types: isolated and non-isolated. BDCs are widely applied due to their simple structure and control. A high switching frequency is a good solution to achieve a high-power density in BDCs. Obviously, hard switching (HS) limits the switching frequency, and increases both the electromagnetic interference (EMI) and switching loss. To solve these problems, soft-switching has been developed in BDCs. The task is the challenging since soft switching must be ensured for the both power flow directions. Studies for soft switching have yielded different topologies, and they can be classified into three basic types.
1) The first type of solution adopts the interleaved topology [7]. In some DC-DC converters, parallel connections are used to form interleaved structures. To realizing soft switching, the charge and discharge of the bypass capacitors of the switches can be buffered by the positive and negative flowing of the inductance current. However, too many components, such as double switches and double main inductors, can result in the power density becoming lower, which makes the control algorithm more complex.
2) The second type of solution is to produce an oscillation between the inductors and capacitors, [9]-[12], which makes it possible for zero voltage or zero current to be obtained. In [13], the auxiliary switch is operated two times in one cycle, which increases the complex of the control method. In [14], the circulating current of the auxiliary switchings are reduced by the conduction time reduction. However, when the auxiliary switch turns off, the oscillation between the inductors and snubber capacitors in series results in an undesirable oscillation, which reduces the converter efficiency and increases the voltage stress of the semiconductor elements. In [15], a complicated soft-switching structure is applied to a converter. However, this has relative drawbacks. In [16], a converter is designed with an auxiliary switch operating twice in a switching period. As a result, the switching loss increases, the control strategy becomes more complex, and the switching frequency is limited. In addition, the main switch suffers from high current stress.
In [17], the bi-directional DC-DC converter has the advantage of fewer components, a simple structure and a simple control method. The boost and buck modes share the same resonant circuits. Therefore, the efficiency is high. However, the resonant inductance used to realize ZVS is located in the main circuit of the topology. Thus, the soft switching characteristics of the topology are affected more by the operating state and load of the main circuit. On the other hand, it leads to instability of the main circuit voltage.
3) The third type of solution uses coupled inductors to provide soft-switching of the auxiliary switches [18]-[24]. The coupled inductor is used as a resonant inductor, and all of the magnetic elements are made on a single magnetic core. Therefore, the converter weight and volume are reduced. However, in some converters, the voltage stress of the auxiliary switch exceeds the high voltage side.
In [25], the authors presented a family of snubber active structures using feedback inductors. Although there are more diodes, the feedback inductance is used to transfer the resonant current to the load side to improve efficiency. However, the coupling factor of the feedback inductor is close to 1. Therefore, the topology cannot control the auxiliary switch current stress or reverse recovery current by designing the coupling coefficient.
In this study, a new ZVZCT bidirectional converter using coupled inductors is introduced. For the proposed converter, the soft switch can be implemented for all of the semiconductor components in the entire duty cycle range, regardless of the power flow direction. For this converter, the main switch is easy to control and does not have extra voltage or current stress. In the boost mode, the main switch turns on with ZVZCT and turns off with ZVS. The auxiliary switch turns on with ZCS and turns off at ZVS. In the buck mode, the main switch can be operated with ZVZCT turn-on and ZVS turn-off, and the auxiliary switch can also realize ZCS turn-on and ZVT turn-off.
For this topology, full coupling of the inductors is not necessary. Because the new topology is able to conduct leakage current through the energy release loop, the parasitic oscillation and current rush are eliminated. The coupling inductance is able to deliver resonant power to the output terminal through the coupling effect, and the current stress of the auxiliary switch can be limited, which reduces the conduction loss and improves the efficiency.
In the boost mode, extra resonance energy is transferred to the load side via the coupled inductor. Therefore, the current stress on the auxiliary switch is controllable. When designing the topology, the freewheeling current can be reduced by adjusting the coupling coefficient. As a result, the design flexibility voltage are low. There is no current or voltage spike, which improves the working conditions of the auxiliary switch. With the same components, the load range for ZVZCT realization is expanded.
The rest of this paper is organized as follows. The second section describes the topology and working principle. The third section provides a parameter analysis and the topology design. In the fourth part, experimental results are discussed. The final section gives some conclusions.
Ⅱ. TOPOLOGY AND WORKING PRINCIPLE
The proposed ZVZCT bidirectional DC-DC converter is shown in Fig. 1(a), and Fig. 1(b) shows an equivalent circuit of this topology. The resonance part includes one resonant capacitor, a coupled-inductor, two diodes and two auxiliary switches, where vL1, vLa1 and vLb1 are the voltages of the inductors; L1, La1 and Lb1 are the inductances; M is the mutual inductance of La1 and Lb1; and iL1, iLa1 and iLb1 are the currents of L1, La1 and Lb1.
|
(a) |
|
(b) |
A. Boost Working Mode
In the boost mode, seven working modes are analyzed in one period. Fig. 2(a) shows each operation mode for the topology. Key waveforms of the boost mode are shown in Fig. 2(b).
|
(a) |
|
(b) |
1) Mode 1 (t0 ≤ t < t1):
This is the main switch off mode for the traditional boost converter. In this interval, both S1 and S2 are off and DS3 is on. VS1 is equal to the output voltage. Therefore, there is no voltage stress on it. The converter energy is transfered from the low side to the high side.
2) Mode 2 (t1 ≤ t < t2):
The initial conditions at t1 are vCS1=VH, vCr2=0, iS1=0, iS2=0, iDS3=Ii and iLa1=iLb1=0. The turn-on signal is implemented to S2, and the resonance begins with La1, Lb1, CS1 and CS3. This resonance provides S2 with ZCS turn-on. Meanwhile, the current iCS3 falls down, and Cr2 is charged. At the end of this mode, the resonance reduces the current of CS3 to zero. Therefore, S3 achieves ZCS turn-off, and the capacitor Cr2 voltage rises. The following equations are obtained in this mode by KVL:
3) Mode 3 (t2 ≤ t < t3):
In mode 3, CS1 and CS3 have been discharged to zero and separately charged to the output voltage. Thus, S3 does not suffer from voltage stress. DS1 is at the turn-on stage, and Cr2 continues the charge. La1, Lb1 and Cr2 resonate though the body diode of S1. Once Cr2 has been completely charged, the voltage of Cr2 is the same as that of C2. Therefore, this condition provides the body diode of S4 ZVS turn-on. At the end of mode 3, the voltage of Cr2 reaches VH. S4 is turned on with ZVS and the current in Lb1 continues. Thus, the energy transfers from the coupled inductor to the load by the body diode of S4 in the next mode. The current stress of S2 is obviously reduced. iLa1= ILa1(t2), iLb1=ILb1(t2) and uLb1=ULb1(t2) are the initial conditions. For this resonance, the following equations are:
(2)Therefore:
(3)
(4)
(5)where 
4) Mode 4 (t3 ≤ t < t4):
In this mode, S1 is turned on. Because the current in Lb1 falls to zero slowly, the body diode of S4 is turned off with ZCS. Meanwhile, the freewheeling current iLa1−Ii in S1 continues to decrease. At t=t3, the initial conditions are vCr2=VH, iLa1=ILa1(t3) and iLb1=ILb1(t3). The following equations are valid:
(7)
The interval before which iLa1−Ii fall to zero provides the ZVZCT turn-on margin for S1. The signal of S1 is applied in this margin when the body diode is active.
5) Mode 5 (t4 ≤ t < t5):
At the beginning of mode 5, S2 is turned off. The turn-off current for S2 is lower than Ii due to the resonance in mode 4. Furthermore, the voltage of S2 is not directly raised to VH. The ZVS-off of S2 is realized due to Cr2. The current in Lb1 and Cr2 resonates. At t=t4, the initial conditions are iLb1=ILb1(t4), iLa1=ILa1(t4), vcr2(t4)=VH, and v=U Lb1(t4). The following equations exist:
(9)Further derivation can be obtained as:
(10)where: 
.
6) Mode 6 (t5 ≤ t < t6):
This is a conventional PWM mode. In this mode, Cr2 is discharged. Therefore, the voltage cross D2, which is parallel to Cr2, also falls to zero, and D2 conducts the current with ZVS. The inductor Lb1 current falls to 0.
7) Mode 7 (t6 ≤ t < t7):
In this mode, the main switch is off and the capacitors CS1 and CS3 are separately charged and discharged. At t=t6, vCS1=0 and vCs3=VH.
In addition, the solution can be obtained by the equation group:
(12)
B. Buck Working Mode
For the buck mode, each operation stage of this topology is presented in Fig. 3(a). Fig. 3(b) shows the key waveforms of the buck mode. The seven operation stages for one PWM period have been analyzed below.
|
(a) |
|
(b) |
1) Mode 1 (t0 < t < t1):
Mode 1 is the PWM standard mode. At t = t0, the main switch S3 is off. L1 and C1 supply the output through the body-diode of S1.
2) Mode 2 (t1 < t < t2):
At the beginning of mode 2, S4 is activated. Resonance starts between Cr2, La1, Lb1, CS3 and CS1. DS1 is turned off with ZVS because of the bypass capacitor CS1. S4 turns on with ZCS due to the coupling inductors. CS1 is charged to VH, and CS3 is discharged. Thus, S1 does not suffer from voltage stress. The initial conditions are vCr2=0, iLa1=iLb1=0, vCs1(t1)=0, vCs3(t1)=VH and IL=constant. The following equations are derived as:
(14)
3) Mode 3 (t2 < t < t3):
In mode 3, resonance separately charges and discharges CS1 and CS3. Later, DS3 is active with ZVS due to the bypass capacitor CS3. Lb1, La1 and Cr2 continue to resonate through the anti-parallel-diode of S3. The following initial values are valid: vCr2= UCr2(t2), iLa1= ILa1(t2), iLb1= ILb1(t2) and uLb1= ULb1(t2). The following formulas can be obtained:
The following conclusions are valid:
(16)where:
.
4) Mode 4 (t3 < t < t4):
For this mode, Cr2 has been completely charged. The current in La1 and Lb1 continues to flow through DS3. t4-t2 are the ZVT interval for S3. At the end of mode 4, the current in La1 and Lb1 decreased to 0. The initial conditions are: vCr2= vCr2(t3) and iLa1= iLb1= ILa1(t3).
From the differential equation, it can be seen that in this mode, iLb1 is close to the constant current ILa1(t3). Then the interval of this mode should not be too long. Otherwise, there are serious circulation losses. This means that once the diode of the auxiliary switch and the main switch are active, the main switch should be turned on as soon as possible.
5) Mode 5 (t4 < t < t5):
When this mode starts, the main switch S3 is turned on. Cr2 starts to discharge, and Cr2 resonates with La1 and Lb1. This resonance provides S4 with ZCS. The active signal is removed when this mode ends. The initial conditions are as follows : vCr2= VH and iLa1= iLb1= 0.
Then:
(19)where: 
6) Mode 6 (t5 < t < t6):
Mode 6 is the PWM standard mode, where VH supplies the output through L1. Meanwhile, Cr2 and Lb1 are oscillating. The initial conditions are as follows: vCr2= UCr2(t5) and iLb1= ILb1(t5). The following equations can be obtained:
Then:
(21)
where: 
7) Mode 7 (t6 < t < t7):
In this mode, Lb1 releases the rest of the energy. At the same time CS1 and CS3 are charged and discharged by the constant current IL1. When this is completed, one circle is over and the next period starts. The initial conditions for this mode are: vCS3=0 and vCS1=VH.
Thus, the solution can be expressed as:
Ⅲ. PARAMETER ANALYSIS AND TOPOLOGY DESIGN
A. ZVZCT Turn-On Conditions and ZVS Turn-Off Conditions for the Main Switch
The ZVS off condition of S1 can always be met due to the bypass capacitor CS1. On the other side, before the activate signal of S2 is off, CS1 must be completely discharged in the boost working mode. At this time, the body diode of S1 starts to conduct current, the driving signal of the main switch (S1) is activated, and the main switch turns on at ZVZCT. In this case, the discharge interval of CS1 and CS3 should not exceed t3-t2. Therefore, the following constraints must be implemented for the ZVZCT turn-on of S1, with consideration of the charge and discharge current of CS1 and CS3:
where Ii is the input current, and iLa1(t3) is the current in La1 for mode 3.
iLa1 creates the ZVS condition for the main switch. When at a light load, the circulating current is larger, while at a heavy load, the circulating current is smaller. The reason for reserving a current margin in the design is to prevent a soft switch lost in the case of iLa1-Ii<0 due to a system overload. This resonant current and the voltage on iS1 do not form an overlap. Thus, there is no power loss for this part.
With the consideration above, the load range can be estimated. If the load changes beyond the designed range, soft switching is lost, the main switch goes into the HS mode, and there is a voltage shock at the main switch. Some circuits that do not work properly in the HS conditions can be damaged.
The turn-on signal of the main switch should delay the auxiliary switch. The delay time should be limited by the conditions below:
where 
, 
is the moment when iDS1, iDS3 decreases to 0, and t3 is the moment when the voltage of CS1 and CS3 resonate to 0.
In the buck mode, to achieve ZVZCT turn-off for S4, the resonance should satisfy the following constraint:
The delay relation between the auxiliary switch and the main switch is related to the load and voltage changes. The main part of the reverse recovery current, ia-Ii, is the key factor in creating the main switch ZVZCT.
Ii is another important factor that determines the delay time. With the consideration of (16) and (27), the following relation can be obtained:
If the main switch voltage is unable to fall back to zero due to factors such as signal mismatch, current chock occurs on the main switch.
The duty cycle D of the conventional buck-boost circuit can affect the soft switching. A D satisfying the following conditions can achieve soft switching:
(31)Obviously, 
The buck mode has a similar conclusion.
If the coupling inductance is not reset, the diode D2 works in the hard switching mode. This forces the inductor to reset when there is still a current flow, which is not desirable. Thus, appropriate circuit parameters are needed to avoid this situation.
The time margin of the reset is tm≈T-ton_aux-trelease>0. Most of the energy in the capacitor Cr2 is transferred to Lb2 and is eventually released through D2. By the equations below, tm can be calculated by:
(33)where, vCr2 (t4) is the voltage after Cr2 is charged, R is the internal resistance of the loop in mode 6, and ton_aux is the pulse width of the auxiliary switch.
B. Coupled Inductor for Soft-Switching Realization
For the proposed topology, the main loop is quite independent. It can adapt to a wide load range. The coupling inductors enable the main switch to acquire soft switching, and assist the auxiliary switch to achieve ZVS or ZCS. The stress on the auxiliary switch can be well controlled by adjusting the coupling coefficient. Therefore, the design of the coupling inductors for this circuit is needed.
Coupled inductors use high frequency iron core materials, such as MnFe204, ZnFe204, etc. In practice, the coupling coefficient of a ring-type core is difficult to reach 0.95. The coupling coefficient varies with the distance between the two windings. Adjustable coupling coefficients can be obtained by adjusting the distance between the two windings.
The coupling coefficient and the values of La1 and Lb1 have a major impact on the circuit. Therefore, in the initial design stage, the magnetic core is an important part. The related parameters are designed as follows.
The initial conditions are: window fill factor: Ko=0.4; core fill factor (ferrite): Kc=1; working flux density of the transformer: Bm≤1/2 𝐵𝑠𝑎𝑡; current density (the value for natural cooling): j=4.2 A/mm2; switching frequency: fsw=100 kHz; output voltage: Vout=200 V; input voltage: Vin=120 V; and auxiliary switch turn-on interval: Ton_max=1 us.
In this design, the OR48X30X15 type is selected. Then Ae = 133 mm2, Aw=1.75 mm2, Le=118 mm and Ve=15700 mm3. In addition, the saturation magnetic density of the ferrite core Bsat = 3900 G. Thus, Bm = 1900 G.
According to simulations: Po=0.000375 W and ILa1_max=12 A. Then:
Due to topology design requirements: La1≤8 uH, it is necessary to take 4 turns. Since the coupling inductance does not want to introduce a higher output voltage (stress), the parameters could be determined according to the features of the circuit topology: Lb1≈La1. Thus, Lb1=6 uH.
(37)AP>APp. The design meets the requirements.
|
(a) |
|
(b) |
The coupling coefficient has a great influence as: 1) the magnitude of the resonant circulation, 2) the realization of the ZVS boundary. It is observed from Fig. 4(a), that once the coupling coefficient exceeds 0.9, S2 and S4 suffer from unacceptable peak currents in both the boost and buck modes. If the coupling coefficient is less than 0.5, the ZVZCT in S4 is lost in the buck mode. Therefore, the best choice is: the maximum current in S2. In addition, S4 and ZVZCT of S4 should both be considered in the buck mode.
Consequently, the following equations become available:
(38)Most of the parameters need to be adjusted in the process of experiments. However, some of the parameters can be estimated by the analysis. Considering that the resonance among CS1, CS3, the coupled inductor and Cr1 occurs in mode 2 of the boost case, Cr2 needs to choose a value with the same level (1-10nF). With equation (15), La1 can be estimated with the assumption of VCr2(t4)=VH (200 V), iLa1(t4)=Ii (the reverse current cannot be too large) and Cr1=5 nF, the relation can be calculated as La1≤8 uH. Therefore, Lb1 should choose the same level (1-10 uH). On the other hand, the coupled inductor has the dotted terminals connected together, and the coefficient k choses the value of 0.8 with consideration of Fig. 4.
With the equation 15, (t3-t2) should be larger than 272 ns with the assumption of Vo=200 V and Ii=5 A. Therefore, with consideration of tdelay > (t3-t1) > (t3-t2), the parasitic capacitance and the switch characteristics, tdelay=1 us. In this coupling inductor design, the value of the coupling inductance is determined first. Then the turn ratio is determined. Therefore, it can meet the soft switching requirement in terms of bidirectional resonance. In addition, the turn ratio of the coupling inductor can also be decided.
C. Switching Loss Distribution
Based on the circuit parameters determined above, it is possible to theoretically estimate the main power distribution of the converter as shown in Fig. 5 below.
|
(a) |
|
(b) |
Thus, the converter loss comes mainly from the switching devices, and the switch loss comes mainly from the turn-on/ turn-off moment. Theoretically, it can be seen that the power cost advantage of soft switching is obvious.
Ⅳ. EXPERIMENTAL RESULTS
In order to verify the above analysis, a prototype of the proposed converter has been established. Table I lists the components and circuit parameters of the converter.
| Symbols |
Descriptions |
Specifications |
| S1, S3 |
Main switch |
IPW60R041C6 |
| S2, S4 |
Auxiliary switch |
IXFK55N50 |
| D2, D3 |
Diodes |
RHRP30120 |
| Cr2 |
Resonant capacitors |
5 nF |
| La1, Lb1 |
Coupled inductors |
8 µH, 6 µH |
| kab |
Coupling coefficients |
0.8 |
A photo of the proposed converter prototype is shown in Fig. 4(b). This converter is composed of a main inductor (L1), the coupled inductors, the capacitor Cr2, and four transistors with body diodes. The output capacitance is 560 uF. The switching frequency of the converter is 100 kHz. The low and high voltages are 120 V and 200 V, respectively. The test power is 500 W.
In the boost mode, control signals are presented in Fig. 6(a), and the processes of the turn-on and turn-off of the main switch are shown in Fig. 6(b). In this figure, vS1 is reduced to zero, and the enable signal is applied to S1. Then the current reverses, and increases from zero. Thus, the S1 ZVZCT turn-on is realized. Once the disable signal is applied, iS1 is rapidly reduced to zero, and vS1 increases. Therefore, S1 also realizes ZVS during the turn-off process.
|
(a) |
|
(b) |
|
(c) |
|
(d) |
The switching process of S2 is shown in Fig. 6(c). The S2 current increases slowly when the active signal of S2 is applied. The coupling inductors are helpful for transferring power to the output side of the converter. Due to the coupling effect, S2 decreases rapidly. Therefore, the smaller average current of S2 reduces the circulating loss when compared with the common inductor. It can also be observed in Fig. 6(c), that when the switching signal of S2 is removed, the ZVS and the lower iS2 and vS2 make S2 have a good turning off.
As shown in Fig. 6(d), the main diode (body diode of S3) in the boost mode operates for soft switching. Therefore, the boost efficiency is improved.
As can be seen in Fig. 7(b), vS3 reduces to “0” before the activation signal of S3 is applied. Then the current of S3 reverses and increases from “0”. Therefore, S3 is able to turn on with ZVZCT. When the signal for S3 is turned off, the zero voltage switch is realized during the turn-off process. In Fig. 7(b), there exists a small overlap in the turn-off, because CS3 only includes the S3 parasitic capacitor. Therefore, the voltage raises quickly. However, this does not get the voltage shock, and it does not significantly increase the power loss. If CS3 increases, this overlap can be avoided. Therefore, choosing a switch component with a larger CS3 may be a solution, while getting an additional parallel capacitor is not recommended.
|
(a) |
|
(b) |
|
(c) |
|
(d) |
The turn-on and turn-off processes for S4 are shown in Fig. 7(c). The current of S4 increases slowly when the active signal of S4 is on. Therefore, S4 turns on with ZCS. Due to the coupling effect, before the signal is removed, the current of S4 resonates to zero and ZVZCT is realized.
In Fig. 7(d) a fly-back diode (body diode of S1) in the buck mode operates under soft switching.
Fig. 8(a) shows that main switch turns on with ZVZCT and turns off with ZVS in the boost mode at a low power level.
|
(a) |
|
(b) |
|
(c) |
In Fig. 8(b), the auxiliary switches turn on with ZCS and turn off with ZVS in the boost mode at a low power level.
As shown in Fig. 8(c), the body diode of S3 is in the soft switching state in the boost mode at a low power level.
It can be seen from Fig. 9(a), that S3 turns on with ZVZCT and turns off with ZVS in the buck mode at a low power level.
|
(a) |
|
(b) |
|
(c) |
The switch S4 in Fig. 9(b) is turned on with ZCS and turned off with ZVZCT in the buck mode at a low power level.
It is shown in Fig. 8(c) that soft switching for the body diode of S1 is obtained for a light load.
The efficiency of the proposed bi-directional converter is tested by a power analyzer (WT-1800) in the boost and buck modes. Efficiency comparison curves for the proposed converter, hard switching and another topology are shown in Fig. 10(a) and (b). These figures show that a high efficiency can be achieved at a full load.
|
(a) |
|
(b) |
At a full load, the peak efficiency reaches 97% and 96.4% in the boost and buck mode for the proposed converter. That is consistent with the fact that the energy in the resonance tank can be delivered to the output and reduce the auxiliary switch average conduction current, due to the coupling inductors.
In [26], the topology is designed for coupling with a main inductor. The resonance inductor is coupled with the main inductor which leads to a large reverse current. Furthermore, an unnecessary oscillation that is not from parasites or stray parameters occurs in the main circuit. This oscillation may cross zero at a low power level. These problems are caused by the resonance and main circuit coupling. For the above reasons, the efficiency at a low power level is not very good.
Another problem with [26], is that the design margin of the coupling inductance parameters is narrow. The soft-switching conditions is depended on a temporary equivalent electromotive force generated by the coupling effect to release the bypass capacitor of the switch. This electromotive force is a function of the main circuit current changing rate, which means that a large ripple must exist in the main circuit to ensure soft switching. Therefore, the ripple rate and soft switch is a pair of contradictions that cannot both occur.
Table II compares a number of topologies including [26]. From this comparison, it is easier to find the features and application scope of each topology.
Ⅴ. CONCLUSION
For this study, a new bi-directional ZVZCT soft-switching converter is proposed. The converter provides the main switch with ZVZCT during the turn-on process and ZVS during the turn-off process, regardless the power flow direction. In addition, the auxiliary switches operate at soft switching due to the coupled inductor. The main diodes can work with soft switching. The main switch does not suffer from additional voltage and current stresses. Furthermore, for the auxiliary switch, the current stress is limited to an acceptable level. A prototype is built, and experimental results verify the realization of the soft switch. This is in accordance with the theoretical analysis derived in this paper.
ACKNOWLEDGMENT
This work was supported by National Science Fund of China (51677118), National Key R&D Plan Key Special Project (2017YFE0102000), Shanghai Municipal Inter-Governmental International Collaboration Project (16510711500) and the International Science & Technology Cooperation Program of China (2016YFE0102200).
REFERENCES
[1] P. Thounthong, “Control of a three-level boost converter based on a differentia flatness approach for fuel cell vehicle applications,” IEEE Trans. Veh. Technol., Vol. 61, No. 3, pp. 1467-1472, Mar. 2012.
[2] C.-E. Kim, S.-K. Han, K.-B. Park, and G.-W. Moon, “A new high efficiency ZVZCS bidirectional dc/dc converter for HEV 42V power systems,” J. Power Electron., Vol. 6, No. 3, pp. 271-278, Jul. 2006.
[3] N. Molavi, E. Adib, and H. Farzanehfard, “Soft-switching bidirectional DC-DC converter with high voltage conversion ratio,” IET Power Electron., Vol. 11, No. 1, pp. 33-42, 2018.
[4] V. V. S. K.Bhajana, P. Drabek, and P. K. Aylapogu, “A novel ZVS non-isolated bidirectional DC-DC converter for energy storage systems,” IECON, pp. 663-668, 2017.
[5] H. Liu, L. Wang, F. Li, and Y. Ji, “Bidirectional active clamp DC-DC converter with high conversion ratio,” Electron. Lett., Vol. 53 No. 22 pp. 1483-1485, 2017.
[6] D. Y. Jung, Y. H. Ji, S. H. Park, Y. C. Jung, and C. Y. Won, “Interleaved soft-switching boost converter for photovoltaic power-generation system,” IEEE Trans. Power Electron., Vol. 26, No. 4, pp. 1137-1145, Apr. 2011.
[7] H. Choi, M. Jang, and V. G. Agelidis, “Zero-current- switching bidirectional interleaved switched-capacitor DC-DC converter: analysis, design and implementation,” IET Power Electron., Vol. 9, No. 5, pp. 1074-1082, 2016.
[8] K. Jin, M. Yang, X. Ruan, S. Member, and M. Xu, “Three- level bidirectional converter for fuel-cell/battery hybrid power system,” IEEE Trans. Ind. Electron., Vol. 57, No. 6, pp. 1976-1986, Jun. 2010.
[9] M. Aamir and H.-J. Kim, “Analysis of ZVS non-isolated bidirectional DC-DC converter,” Circuits and Systems (MWSCAS), pp. 1-4, 2011.
[10] J. K. Eom, J. G. Kim, J. H. Kim, S. T. Oh, Y. C. Jung, and C. Y. Won, “Analysis of a novel soft switching bidirectional DC-DC converter,” J. Power Electron., Vol. 12, No. 6, pp. 859-868, Nov. 2012.
[11] R. N. D. Prado, “A new ZVT PWM converter family: analysis, simulation and experimental results,” APEC, pp. 978-983, 1994.
[12] W. Han and L. Corradini, “Accurate ZVS boundary analysis for bidirectional dual-bridge series resonant DC-DC converters,” Control and Modeling for Power Electronics (COMPEL), 2017.
[13] A. F. Bakan, H. Bodur, and I. Aksoy, “A novel ZVT-ZCT PWM DC-DC converter,” in Proc. 11th Eur. Conf. Power Electron. (EPE), 2005.
[14] P. Das and G. Moschopoulos, “A comparative study of zero-current transition PWM converters,” IEEE Trans. Ind. Electron., Vol. 54, No. 3, pp. 1319-1328, Jun. 2007.
[15] B. Akın and H. Bodur,“ A new single-phase soft-switching power factor correction converter,” IEEE Trans. Power Electron., Vol. 26, No. 2, pp. 436-443, Feb. 2011.
[16] H. Mao, F. C. Y. Lee, X. Zhou, H. Dai, M. Cosan, and D. Boroyevich,“ Improved zero-current transition converters for high-power applications,” IEEE Trans. Ind. Appl., Vol. 33, No. 5, pp. 1220-1232, Sep./Oct. 1997.
[17] P. Das, B. Laan, and S. A. Mousavi, “A nonisolated bidirectional ZVS-PWM active clamped DC-DC converter,” IEEE Trans. Power Electron., Vol. 24, No. 2, pp. 553-558, Feb. 2009.
[18] P. Das, S. A. Mousavi, and G. Moschopoulos, “Analysis and design of a nonisolated bidirectional ZVS-PWM DC-DC converter with coupled inductors,” IEEE Trans. Power Electron., Vol. 25, No. 10, pp. 2630-2641, Oct. 2010.
[19] M. Aamir, S. Mekhilef, and H. J. Kim, “High-gain zero- voltage switching bidirectional converter with a reduced number of switches,” IEEE Trans. Circuits Syst. II: Exp. Briefs, Vol. 62, No. 8, pp. 816-820, Aug. 2015.
[20] Y. T. Chen, S. M. Shiu, and R. H. Liang, “A new family of zero-voltage-transition nonisolated bidirectional converters with simple auxiliary circuit,” IEEE Trans. Ind. Electron., Vol. 63, No. 3, pp.1519-1527, Mar. 2016.
[21] I. Aksoy, H. Bodur, and A. F. Bakan, “A new ZVT-ZCT- PWM DC-DC converter,” IEEE Trans. Power Electron., pp.2093-2105, Aug. 2010.
[22] W. Yu, H. Qian, and J. S. Lai., “Design of high-efficiency bidirectional DC-DC converter and high-precision efficiency measurement,” IEEE Trans. Power Electron., Vol. 25, No. 3, pp. 650-658, Mar. 2010.
[23] G. Chen, Y. Deng, L. Chen, Y. Hu, L. Jiang, X. He, and Y. Wang, “A family of zero-voltage-switching magnetic coupling nonisolated bidirectional DC-DC converters,” IEEE Trans. Ind. Electron., Vol. 64, No. 8, pp. 6223-6233, Aug. 2017.
[24] H. Wu, J. Lu, W. Shi, and Yan Xing, “Nonisolated bidirectional DC-DC converters with negative-coupled inductor,” IEEE Trans. Power Electron., Vol. 27, No. 5, pp. 2231-2235, May 2012.
[25] R. L. Lin, Y. Zhao, and F. C. Lee, “Improved soft-switching ZVT converters with active snubber,” 13th APEC, pp. 1063-1069, 1998.
[26] L. Jiang, C. C. Mi, S. Li, M. Zhang, and X. Zhang, “A novel soft-switching bidirectional DC-DC converter with coupled inductors,” IEEE Trans. Ind. Appl., Vol. 49, No. 6, pp. 2730-2740, Jun. 2013.
Wei Qian received his B.S. degree in Mechanical Engineering and Automation, and his M.S. degree in Aircraft Design from the Nanjing University of Aeronautics and Astronautics (NUAA), Nanjing, China, in 2005 and 2013, respectively. He is presently a Ph.D. candidate in the School of Mechanical Engineering, Shanghai Jiao Tong University (SJTU), Shanghai, China. His current research interests include soft switching power converters, power electronics devices and electric motor control systems for alternative-fuel vehicles.
Xi Zhang received his B.S. degrees in Applied Mathematics and in Information and Control Engineering, and his M.S. and Ph.D. degrees in Power Electronics and Electric Power Drives from Shanghai Jiao Tong University (SJTU), Shanghai, China, in 2002, 2004 and 2007, respectively. From September 2007 to July 2009, he held a postdoctoral position in the Department of Electrical and Computer Engineering, University of Michigan-Dearborn, Dearborn, MI, USA. He is presently working as a Professor in the Institute of Automotive Engineering and in the National Engineering Lab for Automotive Electronics and Control Technology, SJTU. His current research interests include power management strategies, power electronics devices and electric motor control systems for alternative-fuel vehicles.
Zhe Li received his B.E. degree in Mechanical Engineering and Automation from the Department of Mechanical Engineering, Zhengzhou University, Zhengzhou, China, in 2012. He is presently working towards his Ph.D. degree in Mechanical Engineering in the School of Mechanical Engineering, Shanghai Jiao Tong University, Shanghai, China. His current research interests include soft-switching power converters and inductive-based wireless power transfer systems.
Wenqiang Jin received his B.S. degree in Mechanical Engineering from Shanghai Jiao Tong University, Shanghai, China, in 2017, where he is presently working towards his M.S. degree in Vehicle Engineering. Since August 2017, he has been working on project at the Argonne National Laboratory, Lemont, IL, USA. His current research interests include intelligent transportation systems, adaptive traffic signal control, traffic modeling and simulation, and connected vehicle technology.
Jochen Wiedemann received his B.S. degree from Ruhr University Bochum, Bochum, Germany, in 1977; and his M.S. and PhD degrees from the von Karman Institute for Fluid Dynamics, Sint-Genesius-Rode, Belgium, in 1978 and 1983, respectively. He worked at Ruhr University Bochum as a Research Assistant in the Institute of Thermo and Fluid Dynamics from 1977 to 1983. He joined the Department of Bodywork Technology and Aerodynamics at AUDI AG in 1984, where he became a Technical Specialist and a Member of the Extended Management Team in 1991, and a Management-Project Manager for the Audi Wind Tunnel Center in 1995. He has been working as a Professor at the University of Stuttgart, Stuttgart, Germany, as a Chair in the Department of Automotive Engineering, Institute of Internal Combustion Engines and Automotive Engineering, and as a Member of the Board of Directors of the FKFS-Research Institute of Automotive Engineering and Vehicle Engines since 1998. He was a Chairman of the Executive Board of FKFS from 1998 to 2000. He has been teaching Vehicle Aerodynamics at Tongji University, Shanghai, China, since 2002. He has been in charge of a number of research projects including a major DFG instrument program and cooperative projects with nearly all of the world-famous auto enterprises. His current research interests include automotive technology, aerodynamics, driving dynamics, transient aerodynamics, aerodynamics-driving, dynamics interactions, vehicle-driver interaction, acoustics and comfort (NVH), thermal management, etc.