https://doi.org/10.6113/JPE.2019.19.1.58

ISSN(Print): 1598-2092 / ISSN(Online): 2093-4718

Multi-Output LED Driver Integrated with 3-Switch Converter and Passive Current Balance for Portable Applications

Sen Song^*, Kai Ni^*, Guipeng Chen^**, Yihua Hu^*, and Dongsheng Yu^†

^*Department of Electrical Engineering and Electronics, University of Liverpool, Liverpool, ENG, UK

^**School of Aerospace Engineering, Xiamen University, Xiamen, China

^†School of Electrical and Power Engineering, China University of Mining and Technology, Xuzhou, China

Abstract

This study presents a new portable eight-output light emitting diode (LED) driver. The eight output-channels are divided into two equal groups, and their output powers can be controlled individually by three active switches. In addition, a simple capacitor-based passive current balancing circuit (CBC) is employed in each port to guarantee that the currents of the four LEDs are the same. When compared with the conventionally used separate two-output isolated converters, the proposed one uses one less active switch. Moreover, zero-voltage-switching (ZVS) is achieved, which improves the power efficiency of the driver. Finally, a highly compact prototype is built, which can reach an efficiency of 94.6%.

Key words: Multiple outputs, Passive current balance, Reduced switches, Zero-voltage-switching (ZVS)

Manuscript received Apr. 20, 2018; accepted Nov. 10, 2018

Recommended for publication by Associate Editor Fuxin Liu.

^†Corresponding Author: dongsiee@163.com Tel: +86 516 83592000, China Univ. Mining & Tech.

^*Dept. of Electrical Eng. & Electron., University of Liverpool, UK

^**School of Aerospace Engineering, Xiamen University, China

Ⅰ. INTRODUCTION

The light emitting diode (LED) has enormous potential in replacing conventional lamps in residential, automotive, decorative and medical applications due to its advantageous features such as high power density, high luminous efficiency, long lifespan, mercury free nature and quick response [1], [2]. For high illuminance application scenarios, numerous LEDs are connected in series or parallel. For series-connected LEDs, a high voltage stress is caused on the output capacitor and other insulation components. For parallel-connected LEDs, regulating the current through different LED strings requires current balancing technology due to the LED’s current- voltage characteristic and negative temperature coefficient [3].

Achieving a high power efficiency while maintaining high compactness of the whole system is still a challenge for LED drivers. The authors of [4] removed the output capacitor of the buck converter to achieve a high power density, which decreases the power efficiency to under 90%. The authors of [5] used a switched capacitor converter (SCC) to drive LEDs, whose efficiency can only reach around 85%, in spite of the systems simple structure. Fortunately, from the analysis of a SCC [6], [7], significant power losses can be avoided by employing inductors. In [8]-[10], the flyback converter has a simple structure. To increase its power efficiency, different types of snubbers are utilized. However, all of them have their deficiencies. For instance, the active snubber in [8] and the TCR snubber in [9] increase the circuit complexity and cost. The RCD snubber in [10] cannot recycle the leakage inductance power, which leads to hard-switching and causes massive switching losses. With at least two more inductors, the LLC [11], [12] and CLCL topologies [13] can improve the power efficiency by achieving soft-switching operation of the active switches. These projects improve the power efficiency with a sacrifice in terms of compactness, which is not an ideal trade-off.

Current regulating for different LED strings can be achieved by applying active current balancing circuits (CBCs) [14], [15] and passive CBCs [16]-[19]. The active method demands at least one active switch for each output channel, which results in a large switching loss. To avoid high costs and power losses, most current products adopt passive methods, which can be realized by employing inductors [16], [17] or capacitors [18], [19]. The inductor-based method uses transformers to balance the currents through different branches. However, deviations of transformers and the output voltages reduce the balance accuracy. The capacitor-based method can provide precise current balancing with a high power density and a low cost.

This study designs an LED driver with several merits. For instance, it has features such as multi-output, high power efficiency, controllable brightness and reduced component count. With these advantages, the driver is suitable for high power portable applications such as camping lights, vehicle headlights, emergency lights, etc. The topology of the proposed LED driver is shown in Fig. 1.

Fig. 1. Equivalent topology of the proposed LED driver.

This paper is organized as follows. Section II analyses the operation modes of the circuit. Section III discusses some of the main features of the topology. Section IV introduces a design guide for the proposed topology. Section V presents some experiment results. Section VI concludes the paper.

Ⅱ. OPERATION MODE DISCUSSION

The operation principles are discussed in this section. As depicted in Fig. 1, the primary power stage consists of an input DC voltage V_in, three active switches S₀~S₂, two switched capacitors C₁~C₂, and two transformers T₁~T₂ with magnetic inductances L_m1~L_m2 as well as leakage inductances L_k1~L_k2. The three active switches, S₀, S₁ and S₂, control the two output ports: S₀&S₁ for Port 1, and S₀&S₂ for Port 2. The two ports are identical CBCs. More precisely, Port 1 has three resonant capacitors C_r10~Cr₁₂, four diodes D₁₁~D₁₄, four output capacitors C₁₁~C₁₄, and four LED-loads LED₁₁~LED₁₄.

Following assumptions are made to simplify the analysis.

1. All of the switches and diodes are ideal.

2. The transformers T₁ and T₂ are identical, with the same voltage ratio n:1 and secondary side referred leakage inductance L_k.

3. The input voltage is an ideal DC voltage.

4. The capacitance of the resonant capacitors C_r10, C_r11, C_r12, C_r20, C_r21 and C_r22 are equal.
C_r10=C_r11=C_r12=C_r20=C_r21=Cr₂₂=C_r.

5. The voltages of the capacitors C₁~C₂ are constant.

6. The switches S₁ and S₂ have the same duty cycle, D_S1=D_S2=D.

Fig. 2 shows key waveforms corresponding to the eight operating modes depicted in Fig. 3. V_gs0-V_gs2 and V_ds0-V_ds2 are control signals and drain-to-source voltages of the switches S₀-S₂ respectively. V_T1-V_T2 are the primary voltages of the transformers T₁-T₂; V_Cr0-V_Cr2 are the voltages of the three resonant capacitors C_r10-C_r12; and i_D11-i_D14 are the currents through the diodes D₁₁-D₁₄. The two passive CBCs have similar operations. Thus, the operation of Port2 is not discussed.

Fig. 2. Key waveforms of the proposed LED driver.

Fig. 3. Operation Modes 1-8 of the proposed LED driver: (a) Mode 1 [t₀-t₁]; (b) Mode 2 [t₁-t₂]; (c) Mode 3 [t₂-t₃]; (d) Mode 4 [t₃-t₄]; (e) Mode 5 [t₄-t₅]; (f) Mode 6 [t₅-t₆]; (g) Mode 7 [t₆-t₇]; (h) Mode 8 [t₇-t₈].

(a)

(b)

(c)

(d)

(e)

(f)

(g)

(h)

According to the volt-second balance, the voltage across C₁~C₂ can be obtained.

                (1)

                                 (2)

where D = t_on/t_s and t_on are the on-times of the switches S₁~S₂ in each switching cycle.

Mode 1 [t₀–t₁]: At time t₀, the switch S₀ is turned on, and the switch S₁ is off. The voltage across the transformer T₁ is V_in-V_c. The current through L_m1 keeps increasing linearly until t₂.

                        (3)

At time t₀, the total voltage across C_r10, C_r11 and C₁₁ is lower than that of C_r12 and C₁₄.

       (4)

Therefore, D₁₁ is forward biased to transfer power to the LED₁₁. Then C_r10 and C_r11 are charged. The state equations of this mode are obtained as:

               (5)

With Eq. (5), the secondary side current and the voltages across the capacitors C_r10, C_r11 and C_r12 can be calculated as:

       (6)

where Δv_Cr-ini=(V_in-V_C)/n-v_Cr10-v_Cr11-V_C11 is the voltage across the leakage inductance L_k1 at time t₀. Z_n is the characteristic impedance of the resonant tank formed by L_k1, C_r10 and C_r11; and ω_a is the resonant angular frequency. and . At time t₁, the total voltage across C_r10, C_r11 and C₁₁ is the same as that of C_r12.

                  (7)

Then, D₁₄ is conducted. The time duration of Mode 1 is determined as:

            (8)

Mode 2 [t₁–t₂]: From t₁, D₁₄ begins to be forward biased, and the power is transferred to LED₁₁ and LED₁₄. In the secondary circuit, the resonant tank is composed of C_r10, C_r11, C_r12 and L_k1 in this mode. The state equation of Mode 2 is:

       (9)

According to Eq. (9), the secondary current i_s1 is derived as:

     (10)

The time duration of Mode 2 is:

                                        (11)

where t_off is the off-time of the switches S₁ and S₂; k=sinω_aτ₁/sinωbτ_com, θ=π-ωbτ_com is the initial phase of i_s at t₁, is the resonant angular frequency according to Fig. 3(b), and τ_com=t_com-t₁. Where t_com is the time taken for the resonant tank to achieve completing resonance, and t_com can be derived as:

    (12)

Mode 3 [t₂–t₃]: S₂ is turned off at time t₂. During Mode 3, the parasitic capacitor of S₁ is discharged while the parasitic capacitor of S₂ is charged. The voltage across S₁ drops from V_in to zero to achieve ZVS turn-on. The resonant processes of the power stage are given below.

             (13)

where C_s represents the parasitic capacitance of the active switches.

Modes 4-5 [t₃–t₅]: At time t₃, the switch S₁ is turned on with ZVS. During Modes 4-5, the transformer T₁ is powered by the capacitor C₁, and the polarity of the voltage across T₁ is opposite that of Modes 1-3. The current through L_m1 keeps decreasing linearly.

                        (14)

Modes 4-5 are similar to Modes 1-2 except that the three resonant capacitors given the reference of the capacitors here are discharged due to the reversed power flow direction. During Mode 4, D₁₂ is forward biased to transfer power from the input to LED₁₂. At t₄, v_Cr0+v_Cr2+V_C12=v_Cr1+V_C13, and D₁₃ is conducted when power is transferred to LED₁₂ and LED_13.

Mode 6 [t₅–t₆]: S₀ is turned off at t₅. The energy stored in the parasitic capacitor of S₂ is released while the parasitic capacitor of S₀ is charged. The resonant process can be expressed as:

      (15)

Mode 7 [t₆–t₇]: At t₆, the switch S₂ is turned on with zero voltage. Mode 6 takes a relatively short. Thus, it can be neglected. Mode 7 is a continuation of Mode 5.

Mode 8 [t₇–t₈]: S₁ is turned off at t₇. The parasitic capacitor of S₀ is discharged while the parasitic capacitor of S₁ is charged. The voltage across S₀ is decreased with the rising of the voltage across S₁. Therefore, at the start of the next switching cycle, S₀ can achieve ZVS. The resonant process in Mode 8 can be expressed by the following equations:

                     (16)

Ⅲ. TOPOLOGY CHARACTERISTICS ANALYSIS

A. Passive Current Balancing

By the charge balance of the resonant capacitors, the energy charged to C_r11 through D₁₁ in Modes 1-3 is equal to the energy discharged from C_r11 through D₁₃ in Modes 5-8. Because Modes 3, 6 and 8 are very short, they are ignored in this discussion. The balance of C_r11 is between Modes 1-2, 5 and 7. The average current of D₁₁ and D₁₃ in a switching cycle T_s is:

                (17)

where Q_{11_ch} is the energy charged to C_r11 in Modes 1-2, and Q_{11_dis} is the energy discharged from C_r11 in Modes 5 and 7.

Similarly, the charge balance of C_r10 is achieved through D₁₁ and D₁₂. C_r12 is charged through D₁₄ and discharged through D₁₂. Then, Eq. (18)-(20) are obtained as:

               (18)

               (19)

Therefore, according to Eq. (17)-(19), the relationship among the average currents of the diodes D₁₁-D₁₄ are shown as:

              (20)

Additionally, because the output capacitors C_11-14 are large enough, the four output currents and the notations of the four currents are equal to the diodes average currents. Hence:

                     (21)

Due to the CBC, the currents of the four LED-loads are balanced. Moreover, in accordance with Eq. (17)-(20), the non-ideal factors of C_r10, C_r11 and C_r12 are tolerable.

B. Operation of Resonant Capacitors

According to the operations in Modes 1-2 and 4-7, power is delivered to the four output channels through four paths, which are C_r10-C_r11-LED₁₁, C_r12-LED₁₄, C_r10-C_r12-LED₁₂ and C_r11-LED₁₃. When S₁ is open, C_r10-C_r11 and C_r12 are charged. When S₁ is closed, C_r10-C_r12 and C_r11 are discharged. Based on the current balancing analysis, the average currents through LED₁₁ and LED₁₄ are equal. Therefore, Eq. (22) is derived.

    (22)

Hence, at time t₀, Eq. (4) is verified. After t₀, power is delivered to LED₁₁ and then to LED₁₄. Similarly, when S₁ is turned on, power is transferred to LED₁₂ and then to LED₁₃.

C. Control System

As illustrated in Fig. 4(a), the currents through all of the LED strings in Port 1 and Port 2 are controlled by the PWM control where two individual proportion-integrator (PI) compensators are adopted. Due to the current balancing characteristic, the currents through the four LED strings in port 1 LED₁₁-LED₁₄ have the same value as i_o1, and the value of the currents through the four LED strings LED₂₁-LED₂₄ have the same value as i_o2. To collect the output currents i_o1 and i_o2, two sample resistors R_s1 and R_s2 are series connected to one LED string in each port. Thus, i_o1 and i_o2 are regulated as V_o1,ref/R_s1 and V_o2,ref/R_s2. The control variables a and b are received from two individual PI compensators according to the error voltage between the reference voltages V_o1,ref–V_o2,ref and the voltages V_o1 and V_o2 across the sample resistors R_s1 and R_s2.

Fig. 4. Control systems: (a) Control system schematic; (b) Control signals for three switches.

(a)

(b)

As shown in Fig. 4(b), the control signals for the switches S₀, S₁ and S₂ are obtained by comparisons of (a+b, V_t1), (V_t1, a) and (V_t2, b), respectively. D_S0, D_S1 and D_S2 are the duty cycles of the three switches. When compared with the saw tooth V_t1, V_t2 is delayed by a×T_s. To achieve the same operation of two ports, V_o1,ref=V_o2,ref and R_s1 = R_s2 so that the control variables a=b and D_S1=D_S2. The duty cycles of the three switches can be calculated by (23).

                      (23)

To adjust the brightness, the switch S₂ can be short-circuited by an external wire so there is no power transferred to Port 2. The variable c becomes zero. Therefore, the LED driver can provide two-stage brightness: 100% when both S₁ and S₂ are working and 50% when only one of them is operating. Furthermore, the driver provides two-stages brightness and increases reliability due to the two individual PI compensators and two isolated ports.

D. Soft Switching Analysis

Fig. 5. ZVS turn on of three switches in a simulation.

As illustrated in Fig. 5, all three of the power switches can achieve ZVS turn on. In Mode 3, before the switch S₁ is turned on, the energy stored in the parasitic capacitor of the switch S₁ is released to zero. Then the body diode of the switch is forward biased to conduct S₁ with zero voltage. Furthermore, because L_m1 and L_m2 are large enough, the parasitic capacitor of S₁ can be completely discharged. According to Eq. (13), if the dead time t_ds1 is shorter than t₃–t₂, S₁ can achieve the ZVS operation. Similarly, in Eq. (15)-(16), if the dead time t_ds2 is shorter than t₆–t₅, S₂ can obtain ZVS. In addition, S₀ can achieve ZVS when t_ds0<t₈–t₇.

              (24)

where τ₃ is the time duration of mode 3; τ₆ is the time duration of mode 6; and τ₈ is the time duration of mode 8.

E. Expandable Topology

As illustrated in Fig. 6, the converter can be extended in terms of the output channels by adding resonant capacitors. According to the current balancing operation, one phase leg requires one resonant capacitor, e.g. C_r11-C_r12 for phase legs 1-2 respectively, and one more resonant capacitor, e.g. C_r10 is connected between the two phase legs. Hence, by adding 2n-1 resonant capacitors, 2n output channels can be achieved.

Fig. 6. Expandable topology.

Ⅳ. DESIGN GUIDE

This section introduces the design considerations including the relationships among the switching frequency, leakage inductor and the resonant capacitors, as well as the operating characteristics of the components.

A. Passive Current Balancing

The average input current can be calculated as:

          (25)

The output power can be derived as:

          (26)

With the analysis and the trade-off between the power dissipation in the active switches and transformers, the switching frequency is set at 60 kHz in the experiment. To achieve ZVS turn-on of the switches, the on/off time of the switches should be larger than the resonant time based on ω_a and ω_b. As a result, there is always primary current i_p to discharge the parasitic capacitance of the active switches. Moreover, the off-time of the two switches S₁ and S₂ are shorter than their on-time. Hence, according to (12), the following equation can be obtained:

      (27)

where A=arccos{[(V_in-V_C1)/n-v_Cr12(t₀)-V_C14]/Δv_Cr-ini.

However, with a large leakage inductance, the secondary current i_s cannot return to zero immediately after the turn- on/turn-off of active switches. Hence, the leakage inductance is chosen as 9μH, and the 3.3μF resonant capacitance C_r is designed to satisfy Eq. (27).

Based on the primary side referred leakage inductance and switching frequency, the minimum value of the switched capacitors is determined according to:

                            (28)

For the switched capacitors C₁-C₂, a large capacitance can reduce their voltage ripple. However, to reduce the volume of the proposed converter, the switched capacitors are designed as 20μF.

B. Voltage and Current Stress of Active Switches

The voltage stress of three switches is equal to the input voltage V_in. According to Fig. 3, the current through S₁ reaches its maximum value during the interval [t₃~t₅]. With the primary currents, the current stress of the switch S₁ can be obtained as:

       (29)

Due to the large magnetic inductances and the voltage clamping transformers, the currents through the magnetic inductances are ignored. The maximum current of the switch S₂ is obtained at time t₇, which has the same value as that of S₁. The current through S₀ is either i_p1 or i_p2. Therefore, the current stress of the switches can be derived as:

             (30)

where Δv’_Cr-ini=-V_C1/n-v_Cr10+v_Cr12-V_C12 indicates the voltage across the leakage inductance L_k1 at time t₄.

C. Voltage Variation of Resonant Capacitors

According to Eq. (17)-(19) and (22), Eq. (31) can be derived as:

        (31)

where Δv_cr-A and Δv_cr-B are the voltage changes of the corresponding switches in modes 1 and 2, respectively. Additionally, in mode 2, i_Cr12(t)=2i_Cr10(t)=2i_Cr11(t). Hence:

             (32)

where Q_{B_ch} indicates the energy stored in C_r10 or C_r11 in mode 2, and Q_{A_ch} is the energy stored in C_r10 or C_r11 in mode 1. Then the relationships among the voltage and energy variations of three resonant capacitors can be obtained by:

      (33)

Based on Eq. (8), (10) and (33), the following equation is obtained by:

       (34)

With the assumption of C_r10=C_r11=C_r12=C_r, the variables in Eq. (26) can be achieved as θ=1.37rad and k=0.962.

Ⅴ. EXPERIMENT RESULTS

To test and verify the performance of the designed LED driver, a prototype with a 24V DC input is built and displayed in Fig. 7.

Fig. 7. Prototype of the proposed LED driver.

Table I shows the elements employed in the prototype. The power efficiency of this prototype can reach 94.6% with a 60 kHz switching frequency. Due to the similar operation of the two ports, only the experiment results of the primary power stage and Port 1 are illustrated.

TABLE I EXPERIMENT PARAMETERS

Symbols	Definitions	Values
Vin	Input voltage	24 V
Pout	Output power	100 W
Cr	Resonant capacitor	3.3 μF
Cout C1i~C2i	Output capacitor	440μF
C1~C2	Converter capacitor	20μF
Vout	Output voltage	30.8 V
iout	Output current	0.4A
Transformer:
n	Voltage ratio	1:3
Lm	Magnetic inductance	203/201μH
Lk	Leakage inductance	8.27/8.46μH
Core	PQ3525	/
Semiconductor components:
S0~S2	SFP65N06	/
D1i~D2i	MBR20100CT	/

Fig. 8 shows the voltages and primary currents of the two transformers. The obtained experimental waveform is consistent with that in the theoretical analysis. When S₁ is closed, the voltage of the transformer T₁ is negative as –V_c, and the current i_p1 keeps decreasing. When S₁ is open, the voltage across the transformer T₁ is positive as V_in–V_c, and the current i_p1 keeps increasing. Due to the large magnetic inductance, the primary side current i_p1 is mainly affected by the secondary side current i_s1. Moreover, for the same reason, there is a delay for i_p1 to return to zero after the switch’s state changed. Port 2 has a similar operation based on the on/off states of the switch S₂. Additionally, when S₀ is off, the voltages of both transformers are negative.

Fig. 8. Currents and voltages of the transformers T₁~T₂.

As illustrated in Fig. 9, the ZVS operations of the active switches are verified. For example, the drain-source i_ds1 increases to i_p2-i_p1 after S₁ is closed. Meanwhile, when S₀ is open, i_ds1 drops to -i_p1. The drain-source current i_ds flows through the parasitic diode before the turn-on of the corresponding switch and the ZVS turn-on of the switch is achieved. As a result, the switching loss is significantly decreased.

Fig. 9. ZVS operation of three MOSFET switches: (a) S₀; (b) S₁; (c) S₂.

(a)

(b)

(c)

In Fig. 10, the charge and discharge processes of the resonant tank are tested and shown. The two pairs of diodes, D₁₁&D₁₄ and D₁₂&D₁₃ have the same operations. In Fig. 10 (a), D₁₄ is conducted after D₁₁ with a time delay of τ₁. Then they are reverse biased at the same time. Additionally, according to Fig. 10(b), C_r10 and C_r11 are charged before C_r12. Then C_r10 and C_r12 are discharged before C_r11. The voltages across C_r11 and C_r12 are always positive. However, for C_r10, the voltage crosses zero at t₁ and t₄ when the voltages of the two branches are equal. Fig. 10(c) displays the constant balanced output currents of the four output channels in Port 1.

Fig. 10. Current balance operation based on resonant capacitors: (a) currents through the diodes D₁₁-D₁₄; (b) voltages of the resonant capacitors C_r10-C_r12; (c) output currents through the four LED-loads LED₁₁-LED₁₄.

(a)

(b)

(c)

The power efficiency of the LED driver is recorded under different input voltage values as shown in Fig. 11. The red line with diamond markers is obtained when both S₁ and S₂ are working with 8 LED strings. The blue line with triangle markers is obtained when only S₁ is working with 4 LED strings in Port 1. The output power in the case presented by the blue line is half of that of the red line. With 8 LED strings, the efficiency can reach its highest point at 94.6%.

Fig. 11. Power efficiency curve of the proposed LED driver.

Comparative results in terms of design complexity, power efficiency and output power scale are shown in Table II. The non-isolated LED driver with a switched capacitor [5] has the most straightforward design and smallest number of components. The LED driver with a flyback converter [9] uses only one active switch to achieve high compactness. However, their power efficiency is less than 90%, and they are preferred for low power applications. For high illuminance applications, [13] and [16] can achieve a power efficiency of around 95%. However, two external inductors are applied in [13] to construct the CLCL resonant tank, which increases the design complexity and volume. Thanks to the transformers installed for current balancing among different LED strings, [16] is suitable for high power applications, while the non-ideal factors of these transformers result in a high control complexity. From the aforementioned characteristic analysis, the proposed LED driver can achieve a high efficiency of 94.6% with a reduced component count. Additionally, the current balancing of LED strings is obtained by resonant capacitors whose values do not need to be the same.

TABLE II COMPARISON AMONG THE PROPOSED AND OTHER LED DRIVERS

LED drivers	Design complexity	Power efficiency	Output power
Switched capacitor [5]	Low	83%	Low
Flyback converter [9]	Low	90%	Low
CLCL resonant [13]	Medium	94.5%	High
Inductor-based CBC [16]	High	96.4%	High
Proposed	Low	94.6%	High

Ⅵ. CONCLUSION

This study proposes a novel LED driver designed for portable applications. The primary power stage uses only 3 active switches which is one less than conventionally used separate 2 output isolated converters. The following advantages are achieved.

1. High power efficiency: thanks to the ZVS turn-on and the passive CBCs, high power efficiency of 94.6% is obtained.

2. Low design complexity: the power stage uses only two magnetic components and two capacitors to achieve ZVS for all of the active switches. The current balancing operation is achieved due to the three resonant capacitors. Additionally, when compared with the conventional two isolated output converter, the proposed one uses one fewer active switch.

3. Multiple outputs: due to the two separate ports power stage and the two CBCs, the driver provides eight LED output-channels, which are divided into two parts with the same structure.

Experiment results are presented to verify the ZVS of the active switches, the operation of the two isolated ports and the balanced output currents of the proposed LED driver. This project is suitable for high power portable applications such as camping lights, vehicle lights and emergency lights.

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Sen Song was born in Hebei, China. He received his B.S. (with Honors) degrees in Electrical Engineering and Automation from Xi’an Jiaotong-Liverpool University, Suzhou, China, and in Electrical Engineering from the University of Liverpool, Liverpool, ENG, UK, in 2016, where he is presently working towards his Ph.D. degree. His current research interests include power electronic converters and HVDC transmission.

Kai Ni (S’17) was born in Jiangsu, China. He received his B.S. (with Honors) degrees in Electrical Engineering and Automation from Xi’an Jiaotong-Liverpool University, Suzhou, China, and in Electrical Engineering from the University of Liverpool, Liverpool, ENG, UK, in 2016, where he is presently working towards his Ph.D. degree. His current research interests include the operation and control of asynchronized synchronous machines, power electronic converters and power systems.

Guipeng Chen received the B.S. and Ph.D. degrees from the Department of Electrical Engineering, Zhejiang University, Hangzhou, China, in 2011 and 2017, respectively. He joined the Fuji Electric Matsumoto Factory as a Summer Intern in 2014, and was invited to the University of Liverpool, Liverpool, ENG, UK, as a Research Assistant for a half-year program in July 2016. He is presently working as a Postdoctoral Researcher in the Instrument Science and Technology Postdoc Center, School of Aerospace Engineering, Xiamen University, Xiamen, China. His current research interests include automatic topology derivations of dc–dc converters and fault-tolerant converters.

Yihua Hu received his B.S. and Ph.D. degrees from the School of Information and Electrical Engineering, China University of Mining and Technology, Xuzhou, China, in 2003 and 2011, respectively. From 2011 to 2013, he was a Postdoctoral Fellow in the College of Electrical Engineering, Zhejiang University, Hangzhou, China. From 2013 to 2015, he worked as a Research Associate in the Power Electronics and Motor Drive Group, University of Strathclyde, Glasgow, SCT, UK. He is presently working as a Lecturer in the Department of Electrical Engineering and Electronics, University of Liverpool (UoL), Liverpool, ENG, UK. He has published 60 papers in IEEE Transactions journals. His current research interests include renewable generation, power electronics converters and control, electric vehicles, more electric ship/aircraft, smart energy systems and non-destructive test technology. He is the Associate Editor of the IET Renewable Power Generation, IET Intelligent Transport Systems and Power Electronics and Drives.

Dongsheng Yu received his B.S. and Ph.D. degrees from the School of Information and Electrical Engineering, China University of Mining and Technology, Xuzhou, China, in 2005 and 2011, respectively. From 2009 to 2010, he was a Visiting Student at the University of Western Australia, Perth, WA, Australia. From 2014 to 2015, he was an Endeavour Research Fellow at the University of Western Australia. He is presently working as an Associate Professor in the School of Electrical and Power Engineering, China University of Mining and Technology. His current research interests include power electronics, renewable energy, electric drives, nonlinear dynamics and memristive systems. He has published two books and over 50 papers in these areas.