https://doi.org/10.6113/JPE.2018.18.4.1007

ISSN(Print): 1598-2092 / ISSN(Online): 2093-4718

Novel Dual DC-DC Flyback Converter with Leakage-Energy Recycling

Lung-Sheng Yang^†

^†Department of Electrical Engineering, Far East University, Tainan, Taiwan

Abstract

A novel dual DC-DC flyback converter with leakage-energy recycling is presented in this paper. Only an active switch is used for this converter. A pulse-width-modulation strategy is adopeted to control this switch. Two transformers are employed for the proposed converter. During the switch ON-period, the primary windings of the two transformers store energies. At the switch OFF-period, the energies stored in the primary windings of the two transformers are released to the output via the secondary windings of the two transformers. Meanwhile, the leakage energies of the two transformers can be recycled. The operating principles and steady-state analyses of the proposed converter are described in detail. A prototype circuit of the proposed converter is implemented for verifying the performances.

Key words: Flyback converter, Leakage-energy recycling, Pulse width modulation

Manuscript received Oct. 2, 2017; accepted Feb. 26, 2018

Recommended for publication by Associate Editor Yan Xing.

^†Corresponding Author: yanglungsheng@yahoo.com.tw Tel: +886-6-5979566-5410, Far East University

Dept. of Electric Eng., Far East University, Taiwan

Ⅰ. INTRODUCTION

DC power sources are widely utilized in many products, such as communication equipment, medical instruments, and both industrial and commercial devices. It includes nearly all electronics products. Switching mode power converters are applied to the DC power sources. High switching frequencies are used for these converters. Therefore, these converters can provide a high power density, high stability, fast regulation etc. Many topologies, including buck converters [1]-[3], boost converters [4]-[6], buck-boost converters [7], [8], Cuk converters [9], [10] and SEPIC converters [11], [12], have been researched for non-isolated applications. For isolated applications, the forward and flyback converters are used for low-power applications [13]-[18]. In addition, half-bridge, full- bridge and push-pull converters have been utilized for high- power applications [19]-[24]. For isolated and low-power applications, flyback converters are attractive due to their simple structure and low cost. However, the main drawback of flyback converters is the existence of leakage inductance of the transformer. It results in a high voltage spike on the active switch and high power losses. In order to improve this problem, some techniques have been studied. An RCD snubber has been used to limit the voltage spikes on the active switches [25]. However, the energy stored in the leakage inductance of the transformer is dissipated into the resistor of the RCD snubber. A two-switch flyback converter has been used to recycle the leakage-energy of transformers [26], [27]. Nevertheless, this topology need two active switches, which results in higher costs. An active clamp circuit has been applied to eliminate the voltage spike on active switches [28], [29]. In addition, this converter can provide zero voltage switching for the active switch. However, two active switches are employed for this converter. The interleaved flyback converter employs two transformers for high power applications [30]. The leakage energies of the two transformers can be recycled. However, two active switches are employed for this converter as well.

In this paper, a novel single switch dual DC-DC flyback converter with leakage-energy recycling is presented. The circuit configuration of the proposed converter is displayed in Fig. 1. The proposed converter is composed of one switch S₁, two transformers TR₁ and TR₂, three diodes D₁, D₂ and D₃, and three capacitors C₁, C₂ and C_o. The windings in the transformers TR₁ and TR₂ have same turns, N₁₁ = N₂₁ and N₁₂ = N₂₂, and turns ratios n = N₁₂ / N₁₁ = N₂₂ / N₂₁. The leakage energies of the transformers TR₁ and TR₂ can be recycled into the capacitors C₁ and C₂ via the diode D₁ at the switch S₁ OFF-period. Therefore, the power losses can be reduced. For simplifying the circuit analysis of the proposed converter, it is assumed that all of the components are ideal. Therefore, the conducting resistance of the active switch S₁, the forward voltage drop of the diodes D₁, D₂ and D₃, the equivalent series resistance (ESR) of the transformers TR₁ and TR₂ and capacitors C₁, C₂ and C_o are ignored. In addition, these capacitors are sufficiently large. The voltages across these capacitors are considered constant at each switching period.

Fig. 1. Circuit configuration of the proposed converter.

Ⅱ. OPERATING PRINCIPLE

An equivalent circuit of the proposed converter is displayed in Fig. 2. The active switch S₁ is controlled by utilizing the pulse-width modulation strategy. The transformer TR₁ is modeled as the magnetizing inductance L_m11, the leakage inductance L_k11, and an ideal transformer. Similarly, the transformer TR₂ is modeled as the magnetizing inductance L_m21, the leakage inductance L_k21 and an ideal transformer. Because the windings of the transformers TR₁ and TR₂ have same turns, the magnetizing-inductance and leakage- inductance of the transformers TR₁ and TR₂ are assumed as follows:

   (1)

    (2)

Fig. 2. Equivalent circuit of the proposed converter.

The coupling coefficient k of the transformers TR₁ and TR₂ is given as:

       (3)

Some typical waveforms during one switching period in continuous conduction mode (CCM) operation are displayed in Fig. 3. In addition, the equations are assumed as:

   (4)

           (5)

    (6)

   (7)

      (8)

The operating principles in CCM operation are described as follows.

Fig. 3. Some typical waveforms during one switching period of the proposed converter.

Fig. 4. Current direction of the proposed converter: (a) Mode I; (b) Mode II; (c) Mode III; (d) Mode IV.

(a)

(b)

(c)

(d)

Mode I [t₀, t₁]: The active switch S₁ is turned on. The current direction is displayed in Fig. 4(a). The energies stored in the magnetizing inductances L_m11 and L_m21 of the transformers are released to the output capacitor C_o and the load R via ideal transformers and the diodes D₂ and D₃. The DC-source V_in, the capacitor C₁ and the magnetizing inductance L_m21 transfer their energies to the leakage inductance L_k21 in series via the active switch S₁. Similarly, the DC-source V_in, the magnetizing inductance L_m11 and the capacitor C₂ transfer their energies to the leakage inductance L_k11 in series via the active switch S₁. Thus, the magnetizing- inductance currents i_Lm11 and i_Lm21 are decreased and the leakage-inductance currents i_Lk11 and i_Lk21 are increased, as displayed in Fig. 3. This mode ends when the magnetizing- inductance currents i_Lm11 and i_Lm21 are equal to the leakage- inductance currents i_Lk11 and i_Lk21 at the moment t = t₁. The voltages across the magnetizing inductances L_m11 and L_m21 and the leakage inductances L_k11 and L_k21 are found as follows:

        (9)

           (10)

From the above equations, the currents through the magnetizing inductances L_m11 and L_m21 and the leakage inductances L_k11 and L_k21 are derived as:

        (11)

      (12)

Mode II [t₁, t₂]: The active switch S₁ is still turned on. The current direction is displayed in Fig. 4(b). The DC-source V_in and the capacitor C₁ are in series to transfer their energies for the magnetizing inductance L_m21 and the leakage inductance L_k21 via the active switch S₁. Similarly, the DC-source V_in and the capacitor C₂ are in series to transfer their energies for the magnetizing inductance L_m11 and the leakage inductance L_k11 via the active switch S₁. The energy stored in the output capacitor C_o is discharged to the load R. Therefore, the magnetizing-inductance currents i_Lm11 and i_Lm21, and the leakage-inductance currents i_Lk11 are i_Lk21 are increased, as displayed in Fig. 3. This mode ends when the active switch S₁ is turned off at this moment t = t₂. The following equations are given as:

          (13)

   (14)

Owing to:

          (15)

The voltage v_Lk is rewritten as:

     (16)

Substituting (16) into (13) yields:

   (17)

Using (12) and (20), the current i_Lm is derived as:

    (18)

Mode III [t₂, t₃]: The active switch S₁ is turned off. The current direction is displayed in Fig. 4(c). The magnetizing inductance L_m21 and the leakage inductance L_k21 are in series to release their energies to the capacitor C₂ via the diode D₁. Similarly, the magnetizing inductance L_m11 and the leakage inductance L_k11 are series to release their energies to the capacitor C₁ via the diode D₁. Thus, the leakage energies have been recycled. The energies stored in the magnetizing inductances L_m11 and L_m21 are released to the load R via ideal transformers and the diodes D₂ and D₃. The energy of the output capacitor C_o is also discharged to the load R. Therefore, the magnetizing-inductance currents i_Lm11 and i_Lm21 and the leakage-inductance currents i_Lk11 and i_Lk21 are decreased, as displayed in Fig. 3. This mode ends when the leakage- inductance currents i_Lk11 and i_Lk21 are equal to zero at this moment t = t₃. The voltages across the magnetizing inductances L_m11 and L_m21, and the leakage inductances L_k11 and L_k21 are obtained as follows:

        (19)

     (20)

Thus:

        (21)

   (22)

Mode IV [t₃, t₄]: The active switch S₁ is still turned off. The current direction is displayed in Fig. 4(d). The energies stored in the magnetizing inductances L_m11 and L_m21 are released to the output capacitor C_o and the load R via ideal transformers and the diodes D₂ and D₃. Thus, the magnetizing-inductance currents i_Lm11 and i_Lm21 are decreased, as displayed in Fig. 3. This mode ends when the active switch S₁ is turned on at the beginning of the next switching period. The voltages across the magnetizing inductances L_m11 and L_m21 are found as follows:

         (23)

In additon:

        (24)

Ⅲ. STEADY-STATE ANALYSIS

A. Voltage Gain

Since the time durations of mode I are very short when compared to a switching period, mode I is neglected in the following analysis. By using the voltage-second balance principle for the magnetizing inductance L_m11, the following equation is given as:

          (25)

Substituting (17), (19) and (23) into (25), the following equation can be found.

    (26)

If the leakage inductances of the transformers are neglected, the coupling-coefficient k is equal to 1. In additon, the voltage V_c can be written as:

   (27)

Substituting (27) and k = 1 into (26), the voltage gain of the proposed converter is found as follows:

     (28)

From the above equation, it can be seen that the duty ratio D cannot be larger than 0.5. Fig. 5 displays curve of the voltage gain M under different turns ratios.

Fig. 5. Voltage gain of the proposed converter under different turns ratios.

B. Boundary Operating Condition

For the sake of simplicity, the leakage inductance of the transformer ignored. When the proposed converter is operated in the boundary conduction mode, the peak value of the magnetizing-inductance current can be found from (18).

   (29)

The average value of the output-capacitor current can be written as:

       (30)

Using the ampere-second balance principle for the output capacitor, it can be seen that the average value of the output-capacitor current I_co is equal to 0. Therefore:

      (31)

The magnetizing-inductance time constant is defined as:

         (32)

Using (28)-(29) and (31)-(32), the boundary magnetizing- inductance time constant τ_LmB is derived as:

    (33)

A curve of the boundary magnetizing-inductance time constant is plotted in Fig. 6. At , the proposed converter is operated in the CCM.

Fig. 6. Boundary condition of the proposed converter at n = 0.75.

C. Voltage Stresses on the Power Devices

From the operating principle analysis, the voltage stresses on the active switch S₁ and the diodes D₁, D₂, D₃ are given as:

          (34)

         (35)

D. Power Losses Analysis

The average of the input current is written as:

   (36)

During the switch S₁ ON-period, the average and root-mean-square of the switch-current are derived as:

   (37)

          (38)

Thus, the power loss of the switch S₁ is given as follows:

      (39)

where r_S1 is the ON-state resistance of the switch S₁. The average of the capacitor-current are given as:

   (40)

                       (41)

The root-mean-square of the capacitor-current is derived as:

    (42)

                                    (43)

Therefore, the power losses of the capacitors are written as:

   (44)

           (45)

where r_c1 and r_co are the ESRs of the capacitors C₁ and C_o. During the switch S₁ OFF-period, the average of the diode-current is derived as:

       (46)

          (47)

        (48)

Thus, the power losses of the diodes D₁-D₃ in the ON-state forward voltage-drop are found as follows:

    (49)

    (50)

where V_FD1 and V_FD2 are the ON-state forward voltage-drop of the diodes D₁ and D₂. The average of the leakage-inductor currents of the transformers TR₁ and TR₂ are given as:

         (51)

The root-mean-square of the leakage-inductor currents of TR₁ and TR₂ are derived as:

        (52)

Thus, the power losses of the primary-winding of TR₁ and TR₂ are given as:

           (53)

where r₁₁ is the ESR of the primary winding of TR₁. The power losses of the secondary-winding of TR₁ and TR₂ are obtained as:

      (54)

where r₁₂ is the ESR of the secondary winding of TR₁.

Ⅳ. EXPERIMENTAL RESULTS

A prototype circuit has been built in the laboratory to verify the feasibility of the proposed converter. The circuit specifications and components are selected as input voltage V_in = 100 V, output voltage V_o = 48 V, output power P_o = 250 W, switching frequency f_s = 75 kHz, capacitors C₁ = C₂ = 100 μF and C_o = 470 μF, turn-ratio of the transformers n = 0.75, switch S₁ IXTQ52N30P, and diodes D₁, D₂ and D₃: STTH6003CW. Thus, the voltage gain M is equal to 0.48. Substituting the voltage gain M = 0.48 and the turns ratio n = 0.75 into (28), the duty ratio D is found to be 0.28. Substituting the turns ratio n = 0.75 and the duty ratio D = 0.28 into (33), the boundary magnetizing-inductance time constant τ_LmB is derived as 0.92. It is assumed that the proposed converter is operated in the CCM from 40% of the full load. Thus, the load R is 23 Ω. At τ_Lm > τ_LmB, the proposed converter is operated in the CCM. Therefore:

   (55)

The magnetizing-inductance L_m is selected as 285 μH.

Under input voltage V_in = 100 V, output voltage V_o = 48 V and output power P_o = 250 W, some experimental waveforms are shown in Fig. 7. Fig. 7(a) shows waveforms of i_Lk11, i_Lk21 and i_D1. It can be seen that the waveforms i_Lk11 and i_Lk21 are almost same. In addition, the summation of the currents i_Lk11 and i_Lk21 is equal to the current i_D1 during the switch S₁ OFF-period. Thus, it can be ensured that the leakage energies of the transformers can be recycled to the capacitors C₁ and C₂ via the diode D₁. Fig. 7(b) shows the waveforms i_S1, i_N12 and i_N22. It can be seen that the summation of the currents i_Lk11 and i_Lk21 is equal to the current i_S1 during the switch S₁ ON-period. Fig. 7(c) shows waveforms of v_S1, v_D1 and v_D2, which agree with the operating principle and steady-state analysis. Fig. 7(d) shows waveforms of V_c1, V_c2, and V_o in the start-up process. It can be seen that the voltage V_o is controlled at the setting value. Fig. 8 shows experimental waveforms under V_in = 100 V, V_o = 48 V and P_o = 70 W. The load R and the magnetizing-inductance time constant τ_Lm are 33 Ω and 0.648, respectively. Thus, τ_Lm is less than τ_LmB. It can be seen that the proposed converter is operated in the discontinuous conduction mode. Fig. 9 shows the dynamic response of the proposed converter for a load change between 70W and 250W. Fig. 10 shows the measured efficiency of this prototype circuit. The maximum measured efficiency is 94.8% and the measured efficiency is 93.1% under the full-load condition.

Fig. 7. Experimental waveforms under V_in = 100 V, V_o = 48 V and P_o = 250 W: (a) i_Lk11, i_Lk21 and i_D1; (b) i_S1, i_N12 and i_N22; (c) v_S1, v_D1 and v_D2; (d) V_c1, V_c2 and V_o.

(a)

(b)

(c)

(d)

Fig. 8. Experimental waveforms of i_Lk11, i_Lk21 and i_D1 under V_in = 100 V, V_o = 48 V and P_o = 70 W.

Fig. 9. Dynamic response of the proposed converter for a load change between 70W and 250W.

Fig. 10. Measured efficiency at various output powers.

Ⅴ. CONCLUSIONS

The conventional DC-DC flyback converter has the merits of a simple structure and low cost. However, this converter also possesses leakage inductance of the transformer. This results in a lower efficiency. In this paper, only a single switch is used in the proposed converter. This proposed converter employs two transformers with the same inductance. During the switch OFF-period, the energies of the magnetizing-inductance of the transformers are released to the output. Meanwhile, the leakage energies of the transformers can be recycled. From the obtained experimental results, it can be seen that the leakage energies of the transformers have been recycled. In addition, the measured efficiency is 93.1% under the full-load condition and the maximum efficiency is 94.8%.

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Lung-Sheng Yang was born in Taiwan, ROC, in 1967. He received his B.S. degree in Electrical Engineering from the National Taiwan Institute of Technology, Taipei, Taiwan, in 1990; his M.S. degree in Electrical Engineering from the National Tsing-Hua University, Hsinchu, Taiwan, in 1992; and his Ph.D. degree in Electrical Engineering from the National Cheng Kung University, Tainan, Taiwan, in 2007. He is presently working as an Associate Professor in the Department of Electrical Engineering, Far East University, Tainan, Taiwan. His current research interests include power factor correction, dc-dc converters, renewable energy conversion and electronic ballasts.