https://doi.org/10.6113/JPE.2019.19.5.1099
ISSN(Print): 1598-2092 / ISSN(Online): 2093-4718
Analysis of Hybrid Converter with Wide Voltage Range Operation
Bor-Ren Lin
†
†Department of Electrical Engineering, National Yunlin University of Science and Technology, Yunlin, Taiwan
Abstract
A soft switching converter with wide voltage range operation is investigated in this paper. A series resonant converter is implemented to achieve a high circuit efficiency with soft switching characteristics on power switches and rectifier diodes. To improve the weakness of the narrow voltage range in LLC converters, an alternating current (ac) power switch is used on the primary side to select a half-bridge or full-bridge resonant circuit to implement 4:1 voltage range operation. On the secondary- side, another ac power switch is adopted to select a full-wave rectifier or voltage-doubler rectifier to achiever an additional 2:1 output voltage range. Therefore, the proposed resonant converter has the capacity for 8:1 (320V~40V) wide output voltage operation. A single-stage hybrid resonant converter is employed in the study circuit instead of a two-stage dc converter to achiever wide voltage range operation. As a result, the study converter has better converter efficiency. The theoretical analysis and circuit characteristics are verified by experiments with a prototype circuit.
Key words: Hybrid resonant circuit, Soft switching operation, Wide voltage operation
Manuscript received Mar. 8, 2019; accepted May 14, 2019
Recommended for publication by Associate Editor Jongwon Shin.
†Corresponding Author: linbr@yuntech.edu.tw Tel: +886-912312281, Fax: +886-5-5312065
Dept. of Electrical Eng., Nat'l Univ. of Science and Technology, Taiwan
Ⅰ. INTRODUCTION
High power density and high efficiency power converters with wide voltage range operations [1]-[5] have been presented and investigated for the power units in railway vehicles, battery chargers in electric vehicles and outdoor LED lighting systems. On railway system applications, the input voltages of dc converters are from 24V to 110V for the lighting, electric door system, motor drive controller and braking system demands. To meet international standard such as EN50155, the input voltage variation of dc converters must be within ±30% or ±40% of its nominal voltage. For the battery chargers in electric vehicles, the output voltage of dc converters are from 200V to 450V. Dc converters with a variable output voltage and a constant power rating are needed for outdoor LED lighting systems. Therefore, soft switching converters with wide voltage range operation are needed for solar power units, electric vehicles, high speed rail vehicles and outdoor LED lighting systems. Conventional two-stage dc-dc converters [6]-[8] have been used to realize wide voltage operation. The front-stage circuit topologies can be buck, buck-boost or boost circuit topologies and the rear-stage circuits can be full-bridge, half-bridge or push-pull circuits. However, the efficiency of two-stage converters is lower than that of single-stage circuits. Dc-dc converters with series or parallel connections were presented in [9]-[11] to achieve wide input or output voltage range operations. The disadvantage of these topologies is the large number of circuit components, which decreases the efficiency and reliability. Full/half bridge dc-dc circuits with phase-shift control and a wide input/output voltage range were presented in [12] and [13]. However, their control strategy is too complicated to be implemented with general integrated circuits. Resonant circuits have been developed in [14]-[17] to accomplish wide voltage operation. However, the operating voltage range is limited at 4:1, i.e., Vin,max=4Vin,min (or Vo,max=4Vo,min). Dc converters with much wider voltage range operation are normally welcome for railway vehicle power units, solar power conversion and output LED lighting power units.
A hybrid converter with wide output voltage range operation (8:1 output voltage range, 320V~40V) and zero voltage switching is presented and investigated. Full-bridge or half-bridge resonant circuits can be selected on the primary side using an alternating current (ac) power switch to achieve zero-voltage switching characteristic for all of the active devices. Since the fundamental primary-side voltage of a full-bridge circuit is two times the fundamental voltage of a half-bridge circuit, the proposed converter can achieve 4:1 voltage range operation on the input side. A half-bridge voltage-doubler rectifier or a full-bridge diode rectifier can be selected on output side using an auxiliary ac power switch to achieve another 2:1 voltage range operation. Therefore, an 8:1 wide voltage range operation is accomplished in the developed circuit. When compared to conventional two-stage converters or the other wide voltage range converters, the circuit topology and control strategy of the presented hybrid resonant circuit are easy to implement. The theoretical analysis, operation principle and circuit characteristics are investigated and confirmed by experiments based on a laboratory prototype.
Ⅱ. CIRCUIT STRUCTURE
The circuit configuration of the proposed wide voltage range resonant circuit is provided in Fig. 1. S1-S4 are power MOSFETs. Q1 and Q2 are ac power switches implemented by two power MOSFETs with a back-to-back connection. Cr is the resonant capacitor. C1 and C2 are the input split capacitors. Co1 and Co2 are the output split capacitors. Lr is the resonant inductor. T is the isolation transformer. Lm is the magnetizing inductance of T. D1-D4 are the rectifier diodes. DS1-DS4 are the body diodes, and CS1-CS4 are the output capacitances of S1-S4. Vin and Vo are the input and output voltages, and Ro is the output resistor. Lm, Lr and Cr are worked as the LLC resonant tank to realize the advantages of soft switching turn-on characteristic for S1-S4 and turn-off characteristic for D1-D4. Therefore, the switching loss and electromagnetic interference of the proposed converter are decreased and improved. To implement wide voltage operation, two ac switches Q1 and Q2 are used on the primary and secondary sides, respectively. Owing to the on/off states of Q1 and Q2, three output voltage ranges (Vo,min~2Vo,min, 2Vo,min~4Vo,min and 4Vo,min~8Vo,min) are achieved. Fig. 2(a) gives the operating circuit when Vo is on the low voltage range (Vo,min~2Vo,min). For the low voltage range, Q1 is on and Q2 is off. The input side is a half-bridge resonant converter and the output side is a full-bridge diode rectifier. The voltage gain on this equivalent circuit is low when compared to the other two operation ranges. The operating circuit for the medium output voltage range (2Vo,min~4Vo,min) is provided in Fig. 2(b). Q1 and Q2 are both on. The input side is a half-bridge resonant converter and the output side is a half-bridge voltage-doubler rectifier. The voltage gain of this equivalent circuit (Fig. 2(b)) is two times of the gain of the low voltage range operation shown in Fig. 2(a). An equivalent circuit for high voltage range operation is shown in Fig. 2(c). Q1 is off and Q2 is on. The input side is a full-bridge resonant converter and the output side is a voltage-doubler rectifier. The voltage gain of this equivalent circuit (Fig. 2(c)) is two times the gain for medium voltage range operation (Fig. 2(b)).
Fig. 1. Circuit diagram of the proposed converter with wide voltage range operation.
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From the on and off conditions of Q1 and Q2 and the PWM signals of S1-S4, the resonant converter with wide voltage range operation is implement in the proposed converter.
Ⅲ. OPERATION PRINCIPLE
Two ac power switches Q1 and Q2 are controlled for different output voltage ranges. According to the on/off states of Q1 and Q2, the proposed converter has three output voltage operation ranges Vo,min~2Vo,min, 2Vo,min~4Vo,min and 4Vo,min~ 8Vo,min. Theoretical circuit waveforms and equivalent circuits for the three output voltage ranges are provided in Figs. 3~5.
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A. Low Output Voltage Range (Q1 on; S1, S2, Q2 off)
To provide a low voltage output (Fig. 2(a)), Q1 is turned on, and S1, S2 and Q2 are turned off for low voltage output operation. The input side is a half-bridge resonant converter. The output side is a full-wave diode rectifier. The voltage gain at low voltage output operation is GL=2nVo,L/Vin, where n is the turn-ratio of T, and Vo,L is the load voltage under the low output voltage range. Fig. 3 shows theoretical circuit waveforms and equivalent circuits for six operating modes.
Mode 1 [t0, t1]: vCS4 =0 at t0. Owing to iLr(t0)<0, DS4 is forward biased, and S4 is turned on at zero-voltage switching. Cr and Lr are naturally resonant. In mode 1, D1 and D4 conduct so that vLm=nVo, iLm increases, and Co1 and Co2 are charged.
Mode 2 [t1, t2]: If the series resonant frequency fr is greater than the switching frequency, iLr is equal to iLm at t1. Therefore, D1 and D4 are turned off. The components Lm, Cr and Lr are naturally resonant.
Mode 3 [t2, t3]: S4 is turned off at t2. Because of iLr(t2)>0, CS3 (CS4) is discharged (charged). Since iLm(t2)> iLr(t2), D2 and D3 are forward biased, and vLm=-nVo. The capacitances of CS3 and CS4 are about hundreds of a picofarad (pF). Thus, the discharge and charge times in this mode are fast enough, and the current values are almost constant.
Mode 4 [t3, t4]: vCS3=0 at t3. Because of iLr(t3)>0, DS3 is forward biased, and S3 is turned on under zero voltage. In mode 4, D2 and D3 are conducting, vLm=-nVo, and Cr and Lr are naturally resonant.
Mode 5 [t4, t5]: If fr > fsw, the primary current iLr is equal to iLm at t4, and D1-D4 are off. In this mode, Lm, Lr and Cr are naturally resonant.
Mode 6 [t5, Tsw+t0]: S3 is turned off at t5. Due to iLr(t5)<0 and iLm(t5)<iLr(t5), CS3 (CS4) is charged (discharged), and D1 and D4 conduct. At Tsw+t0, vCS4=0. Then, the converter goes to next operating cycle.
B. Medium Output Voltage Range (Q1, Q2 on; S1, S2, D3, D4 off)
Fig. 2(b) provides an equivalent circuit for the medium voltage output. For the medium output voltage range, Q1 and Q2 are on, and S1, S2, D3 and D4 are off. The circuit topology includes a half-bridge converter on the input side and a half-wave voltage-doubler rectifier on the output side. The dc voltage gain of the proposed converter operating at the medium output voltage range is GM=nVo,M/Vin, where Vo,M is the load voltage for medium voltage output operation. Comparing the low and medium output voltage range operations, it can observe that Vo,M=2Vo,L under the same input voltage Vin and dc voltage gains GL=GM. Fig. 4 provides theoretical circuit waveforms and equivalent circuits for six operating modes at the medium output voltage operation.
Mode 1 [t0, t1]: vCS4=0 at t0. Because of iLr(t0)<0, DS4 conducts, and S4 is turned on under zero voltage. Due to iLr > iLm, D1 conducts, Co1 is charged by iD1, and Co2 is discharged to supply the load current. In mode 1, vLm=nVo1=nVo/2, and Lr and Cr are naturally resonant.
Mode 2 [t1, t2]: At t1, iLr=iLm. Thus, D1 is off, and Co1 and Co2 are both discharged to supply the load power. On the primary side, Lm, Cr and Lr are naturally resonant.
Mode 3 [t2, t3]: S4 is turned off at t2. Due to iLr(t2)>0 and iLm(t2)>iLr(t2), CS3 (CS4) is discharged (charged), and D2 conducts. Co1 is discharged, and Co2 is charged by iD2.
Mode 4 [t3, t4]: vCS3=0 at t3. Since iLr(t3)>0, DS3 conducts, and S3 is turned on under zero voltage. On the secondary side, D2 conducts, vLm=-nVo2=-nVo/2 and Co2 is charged. Cr and Lr are naturally resonant in this mode.
Mode 5 [t4, t5]: At t4, iLr=iLm. Thus, D2 is reverse biased, and Co1 and Co2 are discharged. Lm, Lr and Cr are naturally resonant.
Mode 6 [t5, Tsw+t0]: S3 is turned off at t5. Because iLr>iLm and iLr(t5)<0, D1 conducts, and CS3 (CS4) is charged (discharged). At Tsw+t0, vCS4=0. Then, the converter goes to next operating cycle.
C. High Output Voltage Range (Q2 on; Q1, D3, D4 off)
An equivalent circuit for high output voltage range operation is given in Fig. 2(c). Q2 is turned on, and Q1, D3 and D4 are in the off-state. The input side is a full-bridge resonant circuit and the output side is a half-wave voltage-doubler rectifier. The dc voltage gain for high voltage range operation is GH=nVo,H/(2Vin), where Vo,H is the load voltage at high voltage output operation. Comparing GL, GM and GH, it is possible to obtain Vo,H=2Vo,M=4Vo,L under the same dc voltage gains GL=GM=GH. Fig. 5 shows theoretical circuit waveforms and equivalent circuits for six operating modes at high output voltage operation.
Mode 1 [t0, t1]: vCS4=0 at t0. Since iLr(t0)<0, DS1 and DS4 conduct. Therefore, S1 and S4 turn on under zero-voltage switching. D1 conducts, and Co1 is charged. In this mode, vLm=nVo/2, and Cr and Lr are naturally resonant.
Mode 2 [t1, t2]: The primary current iLr is decreased and equal to iLm at t1. Then, the diode D1 becomes reverse biased. In this mode, both Co1 and Co2 are discharged, and Lm, Lr and Cr are naturally resonant.
Mode 3 [t2, t3]: S1 and S4 are turned off at t2. Due to iLr(t2)>0 and iLm(t2)>iLr(t2), CS2 and CS3 discharge. Diode D2 conducts, and Co2 is charged.
Mode 4 [t3, t4]: At time t3, vCS2=vCS3=0. Due to iLr(t3)>0, DS2 and DS3 are forward biased, and S2 and S3 are turned on at zero-voltage switching. In this mode, D2 conducts, Co2 is charged, vLm=-nVo/2, and Cr and Lr are naturally resonant.
Mode 5 [t4, t5]: The primary current iLr is equal iLm at t4. Then, D2 becomes reverse biased. Both Co1 and Co2 are discharged. Lm, Lr and Cr are naturally resonant.
Mode 6 [t5, Tsw+t0]: At time t5, the MOSFETs S2 and S3 are turned off. Due to iLr(t5)>iLm(t5) and iLr(t5)<0, D1 becomes forward biased, and CS1 and CS4 are discharged. At Tsw+t0, vCS1=vCS4=0. Then, the circuit goes to next operating cycle.
Ⅳ. CIRCUIT ANALYSIS
Three voltage ranges can be operated in the proposed circuit by selection the half-bridge or full-bridge resonant circuit on input side and the full-wave rectifier or half-bridge voltage-doubler rectifier on the output side. Since an LLC series resonant circuit is employed on the input side to achieve soft switching operation for both the switches and passive diodes, variable frequency modulation is used to regulate the output voltage. The fundamental harmonic frequency method is selected to analyze and estimate the voltage gains of the resonant tank for three output voltage ranges according to the on/off states of Q1 and Q2. When Q1 is in the on-state under low and medium voltage ranges, a square voltage waveform with voltage values of ±Vin/2 is generated on the voltage vab. If the proposed converter is operated at high voltage range operation, Q1 is off and a square voltage waveform with voltage values of ±Vin is shown on the voltage vab. Therefore, the fundamental value of the root-mean-square (rms) voltage Vab,rms is determined by:
(1)In the same way, the fundamental rms value of the magnetizing voltage VLm,rms is determined by:
(2)Based on the dc output resistor, the ac equivalent resistor is determined by:
(3)Therefore, the resonant tank consists of the rms voltage Vab,rms, Lr, Cr, Lm and Rac. The ac voltage gain of resonant tank is obtained as:
(4)where F = fs/fr is the frequency ratio, Ln = Lm/Lr is the inductor ratio, and 
is quality factor. From (4), the output voltage at different voltage ranges is expressed as:
(5)
(6)
(7)According to the output voltage demand, Q1 and Q2 are controlled to have wide output voltage capability. Since the operating frequency is greater than the series resonant frequency under all of the load conditions and input and output voltage values, the input impedance at the resonant tank is always operated at the inductive impedance. Therefore, the power MOSFETs S1-S4 can be operated under soft switching turn-on.
Ⅴ. DESIGN PROCEDURE AND EXPERIMENTAL RESULTS
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To investigate the effectiveness of the proposed circuit, a laboratory prototype was assembled and experimented on with a constant input voltage Vin=400V which is generated by a power factor correction circuit. The output voltage Vo can be changed from 40V to 320V (8:1 ratio). The design resonant frequency fr is 100kHz. The rated load power Po,max is 400W. Based on the fundamental ac voltage gain in (4), output voltage curves for the three output voltage ranges are provided in Fig. 6 under a 400V input. From (4), it can observe that the ac voltage gains of the resonant tank under the three voltage ranges are identical. In each voltage range, the minimum voltage gain Gmin in (4) is designed at unity, and the maximum voltage gain Gmax is intended at two. Thus, the resonant tank in each voltage range can achieve 2:1 voltage ratio operation, and 8:1 wide voltage ratio operation can be achieved in the proposed converter. To accomplish this function, a low inductor ratio Ln=4.5 and a low quality factor are used. There is a ±2V voltage tolerance in the transient voltages such as 80V (160V) between the low (medium) voltage range and the medium (high) voltage range. Under the low output voltage range, Q1 is on and S1, S2 and Q2 are off. The input side has a half-bridge resonant circuit, and the output side has a full-wave rectifier. The maximum and minimum output voltages are 80V and 40V, respectively. The designed voltage gains at the maximum and minimum output voltages are two and unity, respectively. The turn-ratio n can be calculated in (8).
(8)A TDK ferrite core EER-42 is used for the transformer T, with np=60 turns and ns=12 turns. Under the rated power and Vo=80V conditions, the fundamental equivalent resistance on the primary side approximates:
(9)Based on the given parameters, X=0.2, Ln=4.5 and fr=100kHz, Lr, Lm and Cr are determined in (10)~(12).
(10)
(11)
(12)In the prototype, the selected resonant components are Lr=100mH, Lm=450mH and Cr=25nF. The voltage stresses of the power switches and diodes can be approximately calculated as.
(13)
(14)
(15)
(16)
(17)MOSFETs 48N60DM2 (600V40A) are adopted for Q1 and Q2, 22NM60N (650V16A) are used for S1~S5, and STTH12R06D (600V12A) are adopted for D1~D4. The selected capacitances are C1=C2=300mF/400V and Co1=Co2= 1350mF/400V.
The maximum power of the prototype is 400W. The three output voltage ranges can be controlled in the adopted control strategy. Figs. 7 and 8 give measured results under low output voltage operation with Vo=40V and 78V, respectively. The gating voltages of S1~S4 for Vo=40V are shown in Fig. 7(a). It can be observed that S1 and S2 are off, and that S3 and S4 are turned on or off. The primary side voltages vab and vCr, the and current iLr at full load are given in Fig. 7(b) for Vo=40V. The output diode currents iD1~iD4 are provided in Fig. 7(c) for Vo=40V. It is clear that all of the diodes are turned off under zero current switching. In the same manner, experimental waveforms of the proposed converter for Vo =78V at full load are illustrated in Fig. 8. Comparing the test results in Figs. 7 and 8, the switching frequency at Vo=40V is greater than the switching frequency at Vo=78V. For medium voltage range operation, S1, S2, D3 and D4 are off. Measured results at output voltages of Vo=82V and 158V are provided in Fig. 9 and 10, respectively. Figs. 9(a) and 9(b) provide experimental results of the gating voltages and primary waveforms at Vo=82V. The secondary side currents at Vo=82V are given in Figs. 9(c) and 9(d). Likewise, test results of the primary and secondary side waveforms for an 158V output voltage are demonstrated in Fig. 10. Comparing the test results in Figs. 9 and 10 for medium voltage operation, the converter operated at an 82V output voltage has a higher switching frequency than in 158V output voltage case. For high voltage range operation, Q1, D3 and D4 are off. The input side is a full- bridge series resonant circuit, and the output side is a half- bridge voltage-doubler rectifier. Fig. 11 gives test results under high output voltage operation and Vo=162V. Fig. 12 provides experimental results for Vo=320V and full load conditions. Measured converter efficiencies under a full load are 85.2%, 89.5%, 93.2% and 88.2% at Vo=40V, 82V, 162V and 360V, respectively. For a low voltage (high current) output, high conduction losses are observed on the secondary side. For the 360V output case, the converter has a low switching frequency and a high magnetizing current. Therefore, the primary rms current for Vo=320V is greater than the low output voltage case. Thus, larger conduction losses can be observed on the primary side for a Vo=320V output. Fig. 13(a) shows the step response between Vo=40V and 70V for the low output voltage range and Io=5A. In the same manner, Fig. 13(b) provides the step response between Vo=162V and 320V for the high output voltage range and Io=1A. No serious transient response is observed in the test results.
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Ⅵ. CONCLUSIONS
This paper studies and implements a hybrid resonant converter with wide voltage range operation and low switching losses. To accomplish wide voltage range operation, two additional switches are adopted in a conventional full- bridge series resonant converter. One additional switch is used on the input side to achieve 4:1 output voltage range by the selection of a half-bridge or full-bridge series resonant converter. The other switch is adopted on the output side to realize a full-wave diode rectifier or voltage-doubler rectifier. Therefore, the output voltage range can be further extended to another 2:1 operation range. Since a series resonant converter is adopted in the proposed converter, both the active devices and passive diodes are operated under soft switching. Thus, the switching losses on the power semiconductors can be decreased. Two Schmitt voltage comparators are used in the control scheme to determine three output voltage ranges. There is a time delay when the output voltage is across different output voltage ranges. The theoretical circuit characteristics are confirmed by experiments conducted on a laboratory prototype. Future research will focus on improving the step response speed and avoiding unstable oscillations when the output voltage changes from the low output voltage range to the high output voltage range.
ACKNOWLEDGMENT
This research is supported by the Ministry of Science and Technology, Taiwan, under contract MOST 108-2221-E-224- 022-MY2. The author would like to thank Mr. Ji-Wei Chang for his help to measure the circuit waveforms in the experiment.
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Bor-Ren Lin received his B.S. degree in Electronic Engineering from the National Taiwan University of Science and Technology, Taipei, Taiwan, in 1988; and his M.S. and Ph.D. degrees in Electrical Engineering from the University of Missouri, Columbia, MO, USA, in 1990 and 1993, respectively. From 1991 to 1993, he was a Research Assistant with the Power Electronic Research Center, University of Missouri. Since 1993, he has been with the Department of Electrical Engineering, National Yunlin University of Science and Technology, Yunlin, Taiwan, where he is presently a Distinguished Professor. His current research interests include power-factor correction, multilevel converters, active power filters, and soft-switching converters. Dr. Lin received a Fellowship position from the Institution of Engineering and Technology Association in 2017, and he is an Associate Editor of the Institution of Engineering and Technology Proceedings— Power Electronics. He was a recipient of Research Excellence Awards in 2004, 2005, 2007, 2011 and 2018 from the College of Engineering, National Yunlin University of Science and Technology. He received Best Paper Awards from the 2007 and 2011 IEEE Conference on Industrial Electronics and Applications, the 2007 Taiwan Power Electronics Conference, the 2009 IEEE–Power Electronics and Drive Systems Conference, the 2012 Taiwan Electric Power Engineering Conference, the 2014 IEEE–International Conference on Industrial Technology, and the 2019 IEEE–ICA SYMP Conference.